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Электронный компонент: 5962-8853901EA

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REV. E
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
Precision
Instrumentation Amplifier
AD524
FEATURES
Low Noise: 0.3 V p-p 0.1 Hz to 10 Hz
Low Nonlinearity: 0.003% (G = 1)
High CMRR: 120 dB (G = 1000)
Low Offset Voltage: 50 V
Low Offset Voltage Drift: 0.5 V/ C
Gain Bandwidth Product: 25 MHz
Pin Programmable Gains of 1, 10, 100, 1000
Input Protection, Power OnPower Off
No External Components Required
Internally Compensated
MIL-STD-883B and Chips Available
16-Lead Ceramic DIP and SOIC Packages and
20-Terminal Leadless Chip Carriers Available
Available in Tape and Reel in Accordance
with EIA-481A Standard
Standard Military Drawing Also Available
PRODUCT DESCRIPTION
The AD524 is a precision monolithic instrumentation amplifier
designed for data acquisition applications requiring high accu-
racy under worst-case operating conditions. An outstanding
combination of high linearity, high common mode rejection, low
offset voltage drift and low noise makes the AD524 suitable for
use in many data acquisition systems.
The AD524 has an output offset voltage drift of less than 25
V/
C,
input offset voltage drift of less than 0.5
V/
C, CMR above
90 dB at unity gain (120 dB at G = 1000) and maximum non-
linearity of 0.003% at G = 1. In addition to the outstanding dc
specifications, the AD524 also has a 25 kHz gain bandwidth
product (G = 1000). To make it suitable for high speed data
acquisition systems the AD524 has an output slew rate of 5 V/
s
and settles in 15
s to 0.01% for gains of 1 to 100.
As a complete amplifier the AD524 does not require any exter-
nal components for fixed gains of 1, 10, 100 and 1000. For
other gain settings between 1 and 1000 only a single resistor is
required. The AD524 input is fully protected for both power-on
and power-off fault conditions.
The AD524 IC instrumentation amplifier is available in four
different versions of accuracy and operating temperature range.
The economical "A" grade, the low drift "B" grade and lower
drift, higher linearity "C" grade are specified from 25
C to
+85
C. The "S" grade guarantees performance to specification
over the extended temperature range 55
C to +125
C. Devices
are available in 16-lead ceramic DIP and SOIC packages and a
20-terminal leadless chip carrier.
PRODUCT HIGHLIGHTS
1. The AD524 has guaranteed low offset voltage, offset voltage
drift and low noise for precision high gain applications.
2. The AD524 is functionally complete with pin programmable
gains of 1, 10, 100 and 1000, and single resistor program-
mable for any gain.
3. Input and output offset nulling terminals are provided for
very high precision applications and to minimize offset volt-
age changes in gain ranging applications.
4. The AD524 is input protected for both power-on and power-
off fault conditions.
5. The AD524 offers superior dynamic performance with a gain
bandwidth product of 25 MHz, full power response of 75 kHz
and a settling time of 15
s to 0.01% of a 20 V step (G = 100).
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
Analog Devices, Inc., 1999
FUNCTIONAL BLOCK DIAGRAM
AD524
20k
INPUT
G = 10
+INPUT
G = 100
G = 1000
RG
2
RG
1
4.44k
404
40
PROTECTION
20k
20k
20k
20k
20k
SENSE
REFERENCE
V
b
V
OUT
PROTECTION
AD524A
AD524B
AD524C
AD524S
Model
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Units
GAIN
Gain Equation
(External Resistor Gain
Programming)
Gain Range (Pin Programmable)
1 to 1000
1 to 1000
1 to 1000
1 to 1000
Gain Error
1
G = 1
0.05
0.03
0.02
0.05
%
G = 10
0.25
0.15
0.1
0.25
%
G = 100
0.5
0.35
0.25
0.5
%
G = 1000
2.0
1.0
0.5
2.0
%
Nonlinearity
G = 1
0.01
0.005
0.003
0.01
%
G = 10,100
0.01
0.005
0.003
0.01
%
G = 1000
0.01
0.01
0.01
0.01
%
Gain vs. Temperature
G = 1
5
5
5
5
ppm/
C
G = 10
15
10
10
10
ppm/
C
G = 100
35
25
25
25
ppm/
C
G = 1000
100
50
50
50
ppm/
C
VOLTAGE OFFSET (May be Nulled)
Input Offset Voltage
250
100
50
100
V
vs. Temperature
2
0.75
0.5
2.0
V/
C
Output Offset Voltage
5
3
2.0
3.0
mV
vs. Temperature
100
50
25
50
V/
C
Offset Referred to the
Input vs. Supply
G = 1
70
75
80
75
dB
G = 10
85
95
100
95
dB
G = 100
95
105
110
105
dB
G = 1000
100
110
115
110
dB
INPUT CURRENT
Input Bias Current
50
25
15
50
nA
vs. Temperature
100
100
100
100
pA/
C
Input Offset Current
35
15
10
35
nA
vs. Temperature
100
100
100
100
pA/
C
INPUT
Input Impedance
Differential Resistance
10
9
10
9
10
9
10
9
Differential Capacitance
10
10
10
10
pF
Common-Mode Resistance
10
9
10
9
10
9
10
9
Common-Mode Capacitance
10
10
10
10
pF
Input Voltage Range
Max Differ. Input Linear (V
DL
)
2
10
10
10
10
V
Max Common-Mode Linear (V
CM
)
V
Common-Mode Rejection dc to
60 Hz with 1 k
Source Imbalance
G = 1
70
75
80
70
dB
G = 10
90
95
100
90
dB
G = 100
100
105
110
100
dB
G = 1000
110
115
120
110
dB
OUTPUT RATING
V
OUT
, R
L
= 2 k
10
10
10
10
V
DYNAMIC RESPONSE
Small Signal 3 dB
G = 1
1
1
1
1
MHz
G = 10
400
400
400
400
kHz
G = 100
150
150
150
150
kHz
G = 1000
25
25
25
25
kHz
Slew Rate
5.0
5.0
5.0
5.0
V/
s
Settling Time to 0.01%, 20 V Step
G = 1 to 100
15
15
15
15
s
G = 1000
75
75
75
75
s
NOISE
Voltage Noise, 1 kHz
R.T.I.
7
7
7
7
nV/
Hz
R.T.O.
90
90
90
90
nV
Hz
R.T.I., 0.1 Hz to 10 Hz
G = 1
15
15
15
15
V p-p
G = 10
2
2
2
2
V p-p
G = 100, 1000
0.3
0.3
0.3
0.3
V p-p
Current Noise
0.1 Hz to 10 Hz
60
60
60
60
pA p-p
AD524SPECIFICATIONS
2
REV. E
(@ V
S
= 15 V, R
L
= 2 k
and T
A
= +25 C unless otherwise noted)
40 , 000
R
G
+
1


20 %
12 V
G
2
V
D




12 V
G
2
V
D




12 V
G
2
V
D




12 V
G
2
V
D




40 , 000
R
G
+
1


20 %
40 , 000
R
G
+
1


20 %
40 , 000
R
G
+
1


20 %
AD524A
AD524B
AD524C
AD524S
Model
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Min
Typ
Max
Units
SENSE INPUT
R
IN
20
20
20
20
k
20%
I
IN
15
15
15
15
A
Voltage Range
10
10
10
10
V
Gain to Output
l
l
1
l
%
REFERENCE INPUT
R
IN
40
40
40
40
k
20%
I
IN
15
15
15
15
A
Voltage Range
10
10
10
10
V
Gain to Output
l
1
l
1
%
TEMPERATURE RANGE
Specified Performance
25
+85
25
+85
25
+85
55
+125
C
Storage
65
+150
65
+150
65
+150
65
+150
C
POWER SUPPLY
Power Supply Range
6
15
18
6
15
18
6
15
18
6
15
18
V
Quiescent Current
3.5
5.0
3.5
5.0
3.5
5.0
3.5
5.0
mA
NOTES
1
Does not include effects of external resistor R
G
.
2
V
OL
is the maximum differential input voltage at G = 1 for specified nonlinearity.
V
DL
at the maximum = 10 V/G.
V
D
= Actual differential input voltage.
Example: G = 10, V
D
= 0.50.
V
CM
= 12 V (10/2
0.50 V) = 9.5 V.
Specification subject to change without notice.
All min and max specifications are guaranteed. Specifications shown in boldface are tested on all production units at final electrical test. Results from those tests are used to
calculate outgoing quality levels.
AD524
REV. E
3
AD524
REV. E
4
METALIZATION PHOTOGRAPH
Contact factory for latest dimensions.
Dimensions shown in inches and (mm).
OUTPUT
NULL
G = 10
14
13
12
11
10
9
8
7
6
5
4
3
2
1
16
G = 100 G = 1000
SENSE
OUTPUT
+V
S
REFERENCE
INPUT
NULL
RG
2
+INPUT
INPUT
RG
1
OUTPUT
NULL
INPUT
NULL
V
S
PAD NUMBERS CORRESPOND TO PIN NUMBERS FOR THE
D-16 AND R-16 16-PIN CERAMIC PACKAGES.
0.170 (4.33)
15
0.103
(2.61)
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause permanent
damage to the device. This is a stress rating only; functional operation of the device at
these or any other conditions above those indicated in the operational section of this
specification is not implied. Exposure to absolute maximum rating conditions for
extended periods may affect device reliability.
2
Max input voltage specification refers to maximum voltage to which either input
terminal may be raised with or without device power applied. For example, with
18
volt supplies max V
IN
is
18 volts, with zero supply voltage max V
IN
is
36 volts.
ABSOLUTE MAXIMUM RATINGS
l
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
18 V
Internal Power Dissipation . . . . . . . . . . . . . . . . . . . . . 450 mW
Input Voltage
2
(Either Input Simultaneously) |V
IN
| + |V
S
| . . . . . . . .
<36 V
Output Short Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Storage Temperature Range
(R) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
C to +125
C
(D, E) . . . . . . . . . . . . . . . . . . . . . . . . . . . 65
C to +150
C
Operating Temperature Range
AD524A/B/C . . . . . . . . . . . . . . . . . . . . . . . . 25
C to +85
C
AD524S . . . . . . . . . . . . . . . . . . . . . . . . . . 55
C to +125
C
Lead Temperature (Soldering 60 secs) . . . . . . . . . . . . +300
C
CONNECTION DIAGRAMS
Ceramic (D) and
SOIC (R) Packages
TOP VIEW
(Not to Scale)
16
15
14
13
12
11
10
9
1
2
3
4
5
6
7
8
4
15
5
14
V
S
+V
S
OUTPUT
OFFSET NULL
INPUT
OFFSET NULL
INPUT
+ INPUT
RG
2
INPUT NULL
INPUT NULL
REFERENCE
V
S
+V
S
RG
1
OUTPUT NULL
OUTPUT NULL
G = 10
G = 100
G = 1000
SENSE
OUTPUT
AD524
SHORT TO
RG
2
FOR
DESIRED
GAIN
Leadless Chip Carrier
TOP VIEW
4
5
6
7
8
14
15
16
17
18
1
2
3
20 19
9 10 11 12 13
RG
2
INPUT NULL
NC
INPUT NULL
REFERENCE
+INPUT
INPUT
NC
RG
1
OUTPUT
V
S
+V
S
NC
SENSE
OUTPUT NULL
G = 100
G = 10
SHORT TO
RG
2
FOR
DESIRED
GAIN
OUTPUT
NULL
NC
G = 1000
AD524
NC = NO CONNECT
7
19
5
18
V
S
+V
S
OUTPUT
OFFSET NULL
INPUT
OFFSET NULL
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD524 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
WARNING!
ESD SENSITIVE DEVICE
ORDERING GUIDE
Model
Temperature Ranges
Package Descriptions
Package Options
AD524AD
40
C to +85
C
16-Lead Ceramic DIP
D-16
AD524AE
40
C to +85
C
20-Terminal Leadless Chip Carrier
E-20A
AD524AR-16
40
C to +85
C
16-Lead Gull-Wing SOIC
R-16
AD524AR-16-REEL
40
C to +85
C
Tape & Reel Packaging 13"
AD524AR-16-REEL7
40
C to +85
C
Tape & Reel Packaging 7"
AD524BD
40
C to +85
C
16-Lead Ceramic DIP
D-16
AD524BE
40
C to +85
C
20-Terminal Leadless Chip Carrier
E-20A
AD524CD
40
C to +85
C
16-Lead Ceramic DIP
D-16
AD524SD
55
C to +125
C
16-Lead Ceramic DIP
D-16
AD524SD/883B
55
C to +125
C
16-Lead Ceramic DIP
D-16
5962-8853901EA*
55
C to +125
C
16-Lead Ceramic DIP
D-16
AD524SE/883B
55
C to +125
C
20-Terminal Leadless Chip Carrier
E-20A
AD524SCHIPS
55
C to +125
C
Die
*
Refer to official DESC drawing for tested specifications.
REV. E
5
LOAD RESISTANCE
OUTPUT VOLTAGE SWING V
p-p
30
20
0
10
100
10k
1k
10
Figure 3. Output Voltage Swing vs.
Load Resistance
TEMPERATURE C
INPUT BIAS CURRENT nA
40
30
40
75
125
25
25
75
0
20
30
20
10
10
Figure 6. Input Bias Current vs.
Temperature
FREQUENCY Hz
GAIN V/V
0
10
10M
100
1k
10k 100k
1M
1000
100
10
1
Figure 9. Gain vs. Frequency
SUPPLY VOLTAGE V
OUTPUT VOLTAGE SWING
V
20
15
0
0
5
20
10
15
10
5
Figure 2. Output Voltage Swing vs.
Supply Voltage
SUPPLY VOLTAGE V
INPUT BIAS CURRENT
nA
16
14
0
0
5
20
10
15
8
6
4
2
12
10
Figure 5. Input Bias Current vs.
Supply Voltage
WARM-UP TIME Minutes
V
OS
FROM FINAL VALUE
V
0
1.0
8.0
2.0
3.0
4.0
5.0
6.0
7.0
0
3
4
5
6
1
2
Figure 8. Offset Voltage, RTI, Turn
On Drift
SUPPLY VOLTAGE V
INPUT VOLTAGE
V
20
15
0
0
5
20
10
15
10
5
+25 C
Figure 1. Input Voltage Range vs.
Supply Voltage, G = 1
SUPPLY VOLTAGE V
QUIESCENT CURRENT mA
8.0
6.0
0
0
5
20
10
15
4.0
2.0
Figure 4. Quiescent Current vs.
Supply Voltage
INPUT VOLTAGE V
INPUT BIAS CURRENT
nA
16
14
0
0
5
20
10
15
8
6
4
2
12
10
Figure 7. Input Bias Current vs. Input
Voltage
AD524Typical Characteristics
AD524
REV. E
6
140
0
120
80
60
40
20
100
FREQUENCY Hz
CMRR dB
0
10
10M
100
1k
10k 100k
1M
G = 1000
G = 100
G = 10
G = 1
Figure 10. CMRR vs. Frequency RTI,
Zero to 1k Source Imbalance
FREQUENCY Hz
10
100k
100
1k
10k
140
80
60
40
20
120
100
160
0
+V
S
= 15V dc +
1V p-p SINEWAVE
G = 1000
G = 100
G = 10
POWER SUPPLY REJECTION dB
G = 1
Figure 13. Positive PSRR vs.
Frequency
FREQUENCY Hz
CURRENT NOISE SPECTRAL DENSITY fA/
Hz
100k
10k
0
1
10k
10
100
1k
1000
100
Figure 16. Input Current Noise vs.
Frequency
FREQUENCY Hz
FULL POWER RESPONSE V
p-p
30
20
0
1k
10k
1M
100k
10
G = 1, 10, 100
BANDWIDTH LIMITED
G1000
G100
G10
Figure 11. Large Signal Frequency
Response
FREQUENCY Hz
10
100k
100
1k
10k
140
80
60
40
20
120
100
160
0
G = 1000
G = 10
POWER SUPPLY REJECTION dB
G = 1
G = 100
V
S
= 15V dc +
1V p-p SINEWAVE
Figure 14. Negative PSRR vs.
Frequency
0.1 10Hz
VERTICAL SCALE; 1 DIVISION = 5 V
Figure 17. Low Frequency Noise
G = 1 (System Gain = 1000)
SLEW RATE V/
s
GAIN V/V
10.0
8.0
0
1
1000
10
100
6.0
4.0
2.0
G = 1000
Figure 12. Slew Rate vs. Gain
FREQUENCY Hz
VOLT NSD nV/
Hz
1000
100
0.1
1
10
100k
100
1k
10k
10
1
G = 1
G = 10
G = 100, 1000
G = 1000
Figure 15. RTI Noise Spectral
Density vs. Gain
0.1 10Hz
VERTICAL SCALE; 1 DIVISION = 0.1 V
Figure 18. Low Frequency Noise
G = 1000 (System Gain = 100,000)
AD524
REV. E
7
SETTLING TIME s
0
20
5
10
15
12 TO +12
+4 TO 4
+8 TO 8
+12 TO 12
8 TO +8
4 TO +4
1%
0.1%
0.01%
1%
0.1%
0.01%
OUTPUT
STEP V
Figure 21. Settling Time Gain = 10
Figure 24. Large Signal Pulse
Response and Settling Time
G = 100
Figure 20. Large Signal Pulse
Response and Settling Time G =1
SETTLING TIME s
0
20
5
10
15
12 TO +12
+4 TO 4
+8 TO 8
+12 TO 12
8 TO +8
4 TO +4
OUTPUT
STEP V
1%
0.1%
0.01%
1%
0.1%
0.01%
Figure 23. Settling Time Gain = 100
Figure 26. Large Signal Pulse Re-
sponse and Settling Time G = 1000
SETTLING TIME s
0
20
5
10
15
12 TO +12
+4 TO 4
+8 TO 8
+12 TO 12
8 TO +8
4 TO +4
1%
0.1%
0.01%
1%
0.1%
0.01%
OUTPUT
STEP V
Figure 19. Settling Time Gain = 1
Figure 22. Large Signal Pulse
Response and Settling Time
G = 10
SETTLING TIME s
0
80
20
40
60
12 TO +12
+4 TO 4
+8 TO 8
+12 TO 12
8 TO +8
4 TO +4
OUTPUT
STEP V
1%
0.1%
0.01%
1%
0.1% 0.01%
70
10
30
50
Figure 25. Settling Time Gain = 1000
AD524
REV. E
8
Theory of Operation
The AD524 is a monolithic instrumentation amplifier based on
the classic 3 op amp circuit. The advantage of monolithic con-
struction is the closely matched components that enhance the
performance of the input preamp. The preamp section develops
the programmed gain by the use of feedback concepts. The
programmed gain is developed by varying the value of R
G
(smaller
values increase the gain) while the feedback forces the collector
currents Q1, Q2, Q3 and Q4 to be constant, which impresses
the input voltage across R
G
.
V
S
INPUT
20V p-p
100k
0.1%
+V
S
10k
0.01%
1k
10T
10k
0.1%
AD524
1k
0.1%
100
0.1%
11k
0.1%
G = 10
G = 100
G = 1000
RG
2
RG
1
V
OUT
Figure 27. Settling Time Test Circuit
IN
CH
1
V
B
+V
S
I
2
50 A
I
1
50 A
C4
C3
R53
20k
R54
20k
R52
20k
R55
20k
CH
1
+IN
REFERENCE
SENSE
A3
I
4
50 A
I
3
50 A
CH
2
,
CH
3
, CH
4
R57
20k
R56
20k
A1
A2
RG
1
RG
2
4.44k
404
40
G100
G1000
V
S
V
O
CH
2
, CH
3
,
CH
4
Q2, Q4
Q1, Q3
Figure 28 Simplified Circuit of Amplifier; Gain Is Defined as
((R56 + R57)/(R
G
)) + 1. For a Gain of 1, R
G
Is an Open Circuit
As R
G
is reduced to increase the programmed gain, the trans-
conductance of the input preamp increases to the transconduct-
ance of the input transistors. This has three important advantages.
First, this approach allows the circuit to achieve a very high
open loop gain of 3
10
8
at a programmed gain of 1000, thus
reducing gain-related errors to a negligible 30 ppm. Second, the
gain bandwidth product, which is determined by C3 or C4 and
the input transconductance, reaches 25 MHz. Third, the input
voltage noise reduces to a value determined by the collector
current of the input transistors for an RTI noise of 7 nV/
Hz at
G = 1000.
INPUT PROTECTION
As interface amplifiers for data acquisition systems, instrumen-
tation amplifiers are often subjected to input overloads, i.e.,
voltage levels in excess of the full scale for the selected gain
range. At low gains, 10 or less, the gain resistor acts as a current
limiting element in series with the inputs. At high gains the
lower value of R
G
will not adequately protect the inputs from
excessive currents. Standard practice would be to place series
limiting resistors in each input, but to limit input current to
below 5 mA with a full differential overload (36 V) would re-
quire over 7k of resistance which would add 10 nV
Hz of noise.
To provide both input protection and low noise a special series
protect FET was used.
A unique FET design was used to provide a bidirectional cur-
rent limit, thereby, protecting against both positive and negative
overloads. Under nonoverload conditions, three channels CH
2
,
CH
3
, CH
4
, act as a resistance (
1 k
) in series with the input as
before. During an overload in the positive direction, a fourth
channel, CH
1
, acts as a small resistance (
3 k
) in series with
the gate, which draws only the leakage current, and the FET
limits I
DSS
. When the FET enhances under a negative overload,
the gate current must go through the small FET formed by CH
1
and when this FET goes into saturation, the gate current is
limited and the main FET will go into controlled enhancement.
The bidirectional limiting holds the maximum input current to
3 mA over the 36 V range.
INPUT OFFSET AND OUTPUT OFFSET
Voltage offset specifications are often considered a figure of
merit for instrumentation amplifiers. While initial offset may be
adjusted to zero, shifts in offset voltage due to temperature
variations will cause errors. Intelligent systems can often correct
for this factor with an autozero cycle, but there are many small-
signal high-gain applications that don't have this capability.
+V
s
RG
2
AD712
1/2
9.09k
1k
100
16.2k
1/2
+V
S
V
S
16.2k
1 F
1.62M
1.82k
10
100
1000
1 F
1 F
G1, 10, 100
G1000
V
S
AD524
DUT
Figure 29. Noise Test Circuit
AD524
REV. E
9
Voltage offset and drift comprise two components each; input
and output offset and offset drift. Input offset is that component
of offset that is directly proportional to gain i.e., input offset as
measured at the output at G = 100 is 100 times greater than at
G = 1. Output offset is independent of gain. At low gains, out-
put offset drift is dominant, while at high gains input offset drift
dominates. Therefore, the output offset voltage drift is normally
specified as drift at G = 1 (where input effects are insignificant),
while input offset voltage drift is given by drift specification at a
high gain (where output offset effects are negligible). All input-
related numbers are referred to the input (RTI) which is to say
that the effect on the output is "G" times larger. Voltage offset
vs. power supply is also specified at one or more gain settings
and is also RTI.
By separating these errors, one can evaluate the total error inde-
pendent of the gain setting used. In a given gain configuration
both errors can be combined to give a total error referred to the
input (R.T.I.) or output (R.T.O.) by the following formula:
Total Error R.T.I. = input error + (output error/gain)
Total Error R.T.O. = (Gain
input error) + output error
As an illustration, a typical AD524 might have a +250
V out-
put offset and a 50
V input offset. In a unity gain configura-
tion, the total output offset would be 200
V or the sum of the
two. At a gain of 100, the output offset would be 4.75 mV or:
+250
V + 100(50
V) = 4.75 mV.
The AD524 provides for both input and output offset adjust-
ment. This simplifies very high precision applications and mini-
mize offset voltage changes in switched gain applications. In
such applications the input offset is adjusted first at the highest
programmed gain, then the output offset is adjusted at G = 1.
GAIN
The AD524 has internal high accuracy pretrimmed resistors for
pin programmable gain of 1, 10, 100 and 1000. One of the
preset gains can be selected by pin strapping the appropriate
gain terminal and RG
2
together (for G = 1 RG
2
is not connected).
V
S
+V
S
AD524
G = 10
G = 100
G = 1000
V
OUT
OUTPUT
SIGNAL
COMMON
INPUT
OFFSET
NULL
10k
RG
1
+INPUT
INPUT
RG
2
Figure 30. Operating Connections for G = 100
The AD524 can be configured for gains other than those that
are internally preset; there are two methods to do this. The first
method uses just an external resistor connected between pins 3
and 16, which programs the gain according to the formula
R
G
=
40k
G
=
1
(see Figure 31).
For best results R
G
should be a precision resistor with a low
temperature coefficient. An external R
G
affects both gain accuracy
and gain drift due to the mismatch between it and the internal
thin-film resistors. Gain accuracy is determined by the tolerance
of the external R
G
and the absolute accuracy of the internal resis-
tors (
20%). Gain drift is determined by the mismatch of the
temperature coefficient of R
G
and the temperature coefficient of
the internal resistors ( 50 ppm/
C typ).
40,000
2.105
G =
+1 = 20 20%
V
S
+V
S
AD524
V
OUT
REFERENCE
1k
RG
1
+INPUT
INPUT
RG
2
2.105k
1.5k
Figure 31. Operating Connections for G = 20
The second technique uses the internal resistors in parallel with
an external resistor (Figure 32). This technique minimizes the
gain adjustment range and reduces the effects of temperature
coefficient sensitivity.
40,000
4000
||4444.44
G =
+1 = 20 17%
G = 10
*R|
G = 10
= 4444.44
*R|
G = 100
= 404.04
*R|
G = 1000
= 40.04
*
NOMINAL ( 20%)
V
S
+V
S
AD524
V
OUT
REFERENCE
RG
1
+INPUT
INPUT
RG
2
4k
Figure 32. Operating Connections for G = 20, Low Gain
T.C. Technique
The AD524 may also be configured to provide gain in the out-
put stage. Figure 33 shows an H pad attenuator connected to
the reference and sense lines of the AD524. R1, R2 and R3
should be made as low as possible to minimize the gain variation
and reduction of CMRR. Varying R2 will precisely set the gain
without affecting CMRR. CMRR is determined by the match of
R1 and R3.
RG
2
G = 100
G = 1000
RG
1
R2
5k
R3
2.26k
R
L
R1
2.26k
G =
(R2||40k ) + R1 + R3
(R2||40k )
(R1 + R2 + R3)||R
L
2k
G = 10
V
S
+V
S
AD524
V
OUT
+INPUT
INPUT
Figure 33. Gain of 2000
AD524
REV. E
10
INPUT BIAS CURRENTS
Input bias currents are those currents necessary to bias the input
transistors of a dc amplifier. Bias currents are an additional
source of input error and must be considered in a total error
budget. The bias currents, when multiplied by the source resis-
tance, appear as an offset voltage. What is of concern in calculat-
ing bias current errors is the change in bias current with respect to
signal voltage and temperature. Input offset current is the differ-
ence between the two input bias currents. The effect of offset
current is an input offset voltage whose magnitude is the offset
current times the source impedance imbalance.
V
S
+V
S
AD524
LOAD
TO POWER
SUPPLY
GROUND
a. Transformer Coupled
AD524
LOAD
TO POWER
SUPPLY
GROUND
V
S
+V
S
b. Thermocouple
AD524
LOAD
TO POWER
SUPPLY
GROUND
V
S
+V
S
c. AC Coupled
Figure 34. Indirect Ground Returns for Bias Currents
Table I. Output Gain Resistor Values
Output
Nominal
Gain
R2
R1, R3
Gain
2
5 k
2.26 k
2.02
5
1.05 k
2.05 k
5.01
10
1 k
4.42 k
10.1
Although instrumentation amplifiers have differential inputs,
there must be a return path for the bias currents. If this is not
provided, those currents will charge stray capacitances, causing
the output to drift uncontrollably or to saturate. Therefore,
when amplifying "floating" input sources such as transformers
and thermocouples, as well as ac-coupled sources, there must
still be a dc path from each input to ground.
COMMON-MODE REJECTION
Common-mode rejection is a measure of the change in output
voltage when both inputs are changed equal amounts. These
specifications are usually given for a full-range input voltage
change and a specified source imbalance. "Common-Mode
Rejection Ratio" (CMRR) is a ratio expression while "Common-
Mode Rejection" (CMR) is the logarithm of that ratio. For
example, a CMRR of 10,000 corresponds to a CMR of 80 dB.
In an instrumentation amplifier, ac common-mode rejection is
only as good as the differential phase shift. Degradation of ac
common-mode rejection is caused by unequal drops across
differing track resistances and a differential phase shift due to
varied stray capacitances or cable capacitances. In many appli-
cations shielded cables are used to minimize noise. This tech-
nique can create common mode rejection errors unless the
shield is properly driven. Figures 35 and 36 shows active data
guards that are configured to improve ac common mode rejec-
tion by "bootstrapping" the capacitances of the input cabling,
thus minimizing differential phase shift.
V
OUT
REFERENCE
AD524
V
S
+V
S
100
AD711
G = 100
RG
2
+INPUT
INPUT
Figure 35. Shield Driver, G
100
V
OUT
REFERENCE
AD524
V
S
+V
S
100
AD712
RG
2
+INPUT
INPUT
V
S
RG
1
100
Figure 36. Differential Shield Driver
GROUNDING
Many data acquisition components have two or more ground
pins that are not connected together within the device. These
grounds must be tied together at one point, usually at the sys-
tem power-supply ground. Ideally, a single solid ground would
be desirable. However, since current flows through the ground
wires and etch stripes of the circuit cards, and since these paths
have resistance and inductance, hundreds of millivolts can be
generated between the system ground point and the data
AD524
REV. E
11
acquisition components. Separate ground returns should be
provided to minimize the current flow in the path from the sensi-
tive points to the system ground point. In this way supply currents
and logic-gate return currents are not summed into the same
return path as analog signals where they would cause measure-
ment errors.
Since the output voltage is developed with respect to the poten-
tial on the reference terminal, an instrumentation amplifier can
solve many grounding problems.
0.1
F
0.1
F
DIGITAL P.S.
+5V
C
15V
ANALOG P.S.
1 F
DIG
COM
AD574A
C
+15V
6
OUTPUT
REFERENCE
*
ANALOG
GROUND
AD524
DIGITAL
DATA
OUTPUT
SIGNAL
GROUND
*IF INDEPENDENT; OTHERWISE RETURN AMPLIFIER REFERENCE
TO MECCA AT ANALOG P.S. COMMON
1 F
1 F
0.1
F
0.1
F
AD583
SAMPLE
AND HOLD
Figure 37. Basic Grounding Practice
SENSE TERMINAL
The sense terminal is the feedback point for the instrument
amplifier's output amplifier. Normally it is connected to the
instrument amplifier output. If heavy load currents are to be
drawn through long leads, voltage drops due to current flowing
through lead resistance can cause errors. The sense terminal can
be wired to the instrument amplifier at the load, thus putting
the IxR drops "inside the loop" and virtually eliminating this
error source.
V
V+
X1
AD524
OUTPUT
CURRENT
BOOSTER
(REF)
(SENSE)
R
L
V
IN
+
V
IN
Figure 38. AD524 Instrumentation Amplifier with Output
Current Booster
Typically, IC instrumentation amplifiers are rated for a full
10
volt output swing into 2 k
. In some applications, however, the
need exists to drive more current into heavier loads. Figure 38
shows how a high-current booster may be connected "inside the
loop" of an instrumentation amplifier to provide the required
current boost without significantly degrading overall perfor-
mance. Nonlinearities, offset and gain inaccuracies of the buffer
are minimized by the loop gain of the IA output amplifier. Off-
set drift of the buffer is similarly reduced.
REFERENCE TERMINAL
The reference terminal may be used to offset the output by up
to
10 V. This is useful when the load is "floating" or does not
share a ground with the rest of the system. It also provides a
direct means of injecting a precise offset. It must be remem-
bered that the total output swing is
10 volts to be shared be-
tween signal and reference offset.
When the IA is of the three-amplifier configuration it is neces-
sary that nearly zero impedance be presented to the reference
terminal.
Any significant resistance from the reference terminal to ground
increases the gain of the noninverting signal path, thereby upset-
ting the common-mode rejection of the IA.
In the AD524 a reference source resistance will unbalance the
CMR trim by the ratio of 20 k
/R
REF
. For example, if the refer-
ence source impedance is 1
, CMR will be reduced to 86 dB
(20 k
/1
= 86 dB). An operational amplifier may be used to
provide that low impedance reference point as shown in Figure
39. The input offset voltage characteristics of that amplifier will
add directly to the output offset voltage performance of the
instrumentation amplifier.
V
S
+V
S
AD524
REF
SENSE
LOAD
V
IN
+
V
IN
V
OFFSET
AD711
Figure 39. Use of Reference Terminal to Provide Output
Offset
An instrumentation amplifier can be turned into a voltage-to-
current converter by taking advantage of the sense and reference
terminals as shown in Figure 40.
AD524
REF
SENSE
LOAD
AD711
+INPUT
INPUT
R1
V
X
I
L
V
X
R1
I
L
= =
=
(
1 +
V
IN
R1
)
40,000
R
G
A2
Figure 40. Voltage-to-Current Converter
By establishing a reference at the "low" side of a current setting
resistor, an output current may be defined as a function of input
voltage, gain and the value of that resistor. Since only a small
current is demanded at the input of the buffer amplifier A
2
, the
forced current I
L
will largely flow through the load. Offset and
drift specifications of A
2
must be added to the output offset and
drift specifications of the IA.
AD524
REV. E
12
PROGRAMMABLE GAIN
Figure 41 shows the AD524 being used as a software program-
mable gain amplifier. Gain switching can be accomplished with
mechanical switches such as DIP switches or reed relays. It
should be noted that the "on" resistance of the switch in series
with the internal gain resistor becomes part of the gain equation
and will have an effect on gain accuracy.
The AD524 can also be connected for gain in the output stage.
Figure 42 shows an AD711 used as an active attenuator in the
output amplifier's feedback loop. The active attenuation pre-
sents a very low impedance to the feedback resistors, therefore
minimizing the common-mode rejection ratio degradation.
R2
10k
1 F
35V
V
S
OUTPUT
OFFSET
NULL
+V
S
TO V
AD524
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
20k
20k
20k
404
4.44k
20k
+V
S
20k
20k
40
PROTECTION
PROTECTION
IN
+IN
(+INPUT)
(INPUT)
10k
INPUT
OFFSET
NULL
10pF
20k
AD711
V
S
+V
S
AD7590
V
SS
V
DD
GND
39.2k
28.7k
316k
1k
1k
1k
V
DD
A2 A3 A4
WR
V
OUT
Figure 42. Programmable Output Gain
Y0
Y2
Y1
+5V
INPUT
OFFSET
TRIM
C1
C2
ANALOG
COMMON
A
B
INPUTS
GAIN
RANGE
LOGIC
COMMON
+5V
G = 10
K1
G = 100
K2
G = 1000
K3
RELAY
SHIELDS
IN
+IN
V
S
K1 K3 =
THERMOSEN DM2C
4.5V COIL
D1 D3 = IN4148
OUT
K1
K2
K3
D1
D2
D3
10 F
NC
GAIN TABLE
A
B GAIN
0
0
1
1
0
1
0
1
10
1000
100
1
OUTPUT
OFFSET
TRIM
R2
10k
+V
S
74LS138
DECODER
7407N
BUFFER
DRIVER
1 F
35V
NC = NO CONNECT
R1
10k
A1
AD524
1
2
3
4
5
6
7
8
20k
20k
20k
404
4.44k
20k
+V
S
20k
20k
40
PROTECTION
PROTECTION
16
15
14
13
12
11
10
9
Figure 41. Three Decade Gain Programmable Amplifier
AD524
REV. E
13
16
6
+V
S
WR
14
7
15
2
V
OUT
1
18
19
3
17
10
6
2
20
1
4
V
b
AD524
DAC A
DB0
256:1
1/2
AD712
20k
+INPUT
(INPUT)
G = 10
INPUT
(+INPUT)
G = 100
G = 1000
RG
2
RG
1
4.44k
404
40
PROTECTION
20k
20k
20k
20k
20k
DAC B
DB7
AD7528
5
DATA
INPUTS
CS
DAC A
/DAC B
1/2
AD712
9
16
11
12
PROTECTION
3
13
Figure 43. Programmable Output Gain Using a DAC
Another method for developing the switching scheme is to use a
DAC. The AD7528 dual DAC, which acts essentially as a pair
of switched resistive attenuators having high analog linearity and
symmetrical bipolar transmission, is ideal in this application.
The multiplying DAC's advantage is that it can handle inputs of
either polarity or zero without affecting the programmed gain.
The circuit shown uses an AD7528 to set the gain (DAC A) and
to perform a fine adjustment (DAC B).
AUTOZERO CIRCUITS
In many applications it is necessary to provide very accurate
data in high gain configurations. At room temperature the offset
effects can be nulled by the use of offset trimpots. Over the
operating temperature range, however, offset nulling becomes a
problem. The circuit of Figure 44 show a CMOS DAC operat-
ing in the bipolar mode and connected to the reference terminal
to provide software controllable offset adjustments.
WR
CS
+INPUT
G = 10
INPUT
G = 100
G = 1000
RG
2
RG
1
+V
S
AD7524
+V
S
V
S
V
S
+V
S
OUT2
39k
AD589
MSB
LSB
DATA
INPUTS
V
S
1/2
AD712
1/2
AD712
R3
20k
R4
10k
R6
5k
C1
GND
R5
20k
OUT1
V
REF
AD524
Figure 44. Software Controllable Offset
In many applications complex software algorithms for autozero
applications are not available. For those applications Figure 45
provides a hardware solution.
RG
2
RG
1
+V
S
V
S
8
AD524
15
16
14
13
V
DD
GND
0.1 F LOW
LEAKAGE
10
CH
1k
V
OUT
9
10
A1
A2
A3
A4
200 s
V
SS
ZERO PULSE
AD7510KD
AD711
11
12
Figure 45. Autozero Circuit
AD524
REV. E
14
Table II. Error Budget Analysis of AD524CD in Bridge Application
Effect on
Effect on
Absolute
Absolute
Effect
AD524C
Accuracy
Accuracy
on
Error Source
Specifications
Calculation
at T
A
= +25 C at T
A
= +85 C
Resolution
Gain Error
0.25%
0.25% = 2500 ppm
2500 ppm
2500 ppm
Gain Instability
25 ppm
(25 ppm/
C)(60
C) = 1500 ppm
1500 ppm
Gain Nonlinearity
0.003%
0.003% = 30 ppm
30 ppm
Input Offset Voltage
50
V, RTI
50
V/20 mV =
2500 ppm
2500 ppm
2500 ppm
Input Offset Voltage Drift
0.5
V/
C
(
0.5
V/
C)(60
C) = 30
V
30
V/20 mV = 1500 ppm
1500 ppm
Output Offset Voltage*
2.0 mV
2.0 mV/20 mV = 1000 ppm
1000 ppm
1000 ppm
Output Offset Voltage Drift*
25
V/
C
(
25
V/
C)(60
C)= 1500
V
1500
V/20 mV = 750 ppm
750 ppm
Bias Current-Source
15 nA
(
15 nA)(100
) = 1.5
V
Imbalance Error
1.5
V/20 mV = 75 ppm
75 ppm
75 ppm
Bias Current-Source
100 pA/
C
(
100 pA/
C)(100
)(60
C) = 0.6
V
Imbalance Drift
0.6
V/20 mV= 30 ppm
30 ppm
Offset Current-Source
10 nA
(
10 nA)(100
) = 1
V
Imbalance Error
1
V/20 mV = 50 ppm
50 ppm
50 ppm
Offset Current-Source
100 pA/
C
(100 pA/
C)(100
)(60
C) = 0.6
V
Imbalance Drift
0.6
V/20 mV = 30 ppm
30 ppm
Offset Current-Source
10 nA
(10 nA)(175
) = 3.5
V
Resistance-Error
3.5
V/20 mV = 87.5 ppm
87.5 ppm
87.5 ppm
Offset Current-Source
100 pA/
C
(100 pA/
C)(175
)(60
C) = 1
V
Resistance-Drift
1
V/20 mV = 50 ppm
50 ppm
Common Mode Rejection
115 dB
115 dB = 1.8 ppm
5 V = 8.8
V
5 V dc
8.8
V/20 mV = 444 ppm
444 ppm
444 ppm
Noise, RTI
(0.1 Hz10 Hz)
0.3
V p-p
0.3
V p-p/20 mV = 15 ppm
15 ppm
Total Error
6656.5 ppm
10516.5 ppm
45 ppm
*Output offset voltage and output offset voltage drift are given as RTI figures.
AD524C
RG
2
RG
1
+V
S
V
S
G = 100
10k
+10V
350
350
350
350
14-BIT
ADC
0V TO 2V
F.S.
Figure 46. Typical Bridge Application
ERROR BUDGET ANALYSIS
To illustrate how instrumentation amplifier specifications are
applied, we will now examine a typical case where an AD524 is
required to amplify the output of an unbalanced transducer.
Figure 46 shows a differential transducer, unbalanced by 100
,
supplying a 0 to 20 mV signal to an AD524C. The output of the
IA feeds a 14-bit A-to-D converter with a 0 to 2 volt input volt-
age range. The operating temperature range is 25
C to +85
C.
Therefore, the largest change in temperature
T within the
operating range is from ambient to +85
C (85
C 25
C = 60
C).
In many applications, differential linearity and resolution are of
prime importance. This would be so in cases where the absolute
value of a variable is less important than changes in value. In
these applications, only the irreducible errors (45 ppm = 0.004%)
are significant. Furthermore, if a system has an intelligent pro-
cessor monitoring the A-to-D output, the addition of a auto-
gain/autozero cycle will remove all reducible errors and may
eliminate the requirement for initial calibration. This will also
reduce errors to 0.004%.
AD524
REV. E
15
Figure 47 shows a simple application, in which the variation of
the cold-junction voltage of a Type J thermocouple-iron(+)
constantanis compensated for by a voltage developed in series
by the temperature-sensitive output current of an AD590 semi-
conductor temperature sensor.
+V
S
V
S
AD524
V
A
T
A
I
A
IRON
MEASURING
JUNCTION
CONSTANTAN
AD590
V
T
CU
R
A
52.3
R
T
8.66k
1k
E
O
2.5V
AD580
7.5V
+V
S
G = 100
OUTPUT
AMPLIFIER
OR METER
NOMINAL VALUE
9135
E
O
= V
T
V
A
+
2.5V
52.3 I
A
+ 2.5V
1 +
52.3
R
V
T
TYPE
R
A
NOMINAL
VALUE
52.3
41.2
61.4
40.2
5.76
J
K
E
T
S, R
REFERENCE
JUNCTION
+15 C < T
A
< +35 C
Figure 47. Cold-Junction Compensation
The circuit is calibrated by adjusting R
T
for proper output voltage
with the measuring junction at a known reference temperature
and the circuit near 25
C. If resistors with low tempcos are
used, compensation accuracy will be to within
0.5
C, for
temperatures between +15
C and +35
C. Other thermocouple
types may be accommodated with the standard resistance values
shown in the table. For other ranges of ambient temperature,
the equation in the figure may be solved for the optimum values
of R
T
and R
A
.
The microprocessor controlled data acquisition system shown in
Figure 48 includes both autozero and autogain capability. By
dedicating two of the differential inputs, one to ground and one
to the A/D reference, the proper program calibration cycles can
eliminate both initial accuracy errors and accuracy errors over
temperature. The autozero cycle, in this application, converts a
number that appears to be ground and then writes that same
number (8-bit) to the AD7524, which eliminates the zero error
since its output has an inverted scale. The autogain cycle con-
verts the A/D reference and compares it with full scale. A multi-
plicative correction factor is then computed and applied to
subsequent readings.
For a comprehensive study of instrumentation amplifier design
and applications, refer to the Instrumentation Amplifier Applica-
tion Guide,
available free from Analog Devices.
V
REF
AD524
AD7524
AD574A
AD583
AGND
V
REF
V
I N
ADDRESS BUS
20k
20k
10k
5k
AD7507
EN A1
A0 A2
1/2
AD712
RG
2
RG
1
CONTROL
DECODE
LATCH
ADDRESS BUS
1/2
AD712
MICRO-
PROCESSOR
Figure 48. Microprocessor Controlled Data Acquisition System
AD524
REV. E
16
PRINTED IN U.S.A.
C722e04/99
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
20-Terminal Leadless Chip Carrier
(E-20A)
TOP
VIEW
0.358 (9.09)
0.342 (8.69)
SQ
1
20
4
9
8
13
19
BOTTOM
VIEW
14
3
18
0.028 (0.71)
0.022 (0.56)
45
TYP
0.015 (0.38)
MIN
0.055 (1.40)
0.045 (1.14)
0.050 (1.27)
BSC
0.075 (1.91)
REF
0.011 (0.28)
0.007 (0.18)
R TYP
0.095 (2.41)
0.075 (1.90)
0.100 (2.54) BSC
0.200 (5.08)
BSC
0.150 (3.81)
BSC
0.075
(1.91)
REF
0.358
(9.09)
MAX
SQ
0.100 (2.54)
0.064 (1.63)
0.088 (2.24)
0.054 (1.37)
16-Lead Ceramic DIP
(D-16)
16
1
8
9
0.080 (2.03) MAX
0.310 (7.87)
0.220 (5.59)
PIN 1
0.005 (0.13) MIN
0.100
(2.54)
BSC
SEATING
PLANE
0.023 (0.58)
0.014 (0.36)
0.060 (1.52)
0.015 (0.38)
0.200 (5.08)
MAX
0.200 (5.08)
0.125 (3.18)
0.070 (1.78)
0.030 (0.76)
0.150
(3.81)
MAX
0.840 (21.34) MAX
0.320 (8.13)
0.290 (7.37)
0.015 (0.38)
0.008 (0.20)
16-Lead SOIC
(R-16)
16
9
8
1
0.4133 (10.50)
0.3977 (10.00)
0.4193 (10.65)
0.3937 (10.00)
0.2992 (7.60)
0.2914 (7.40)
PIN 1
SEATING
PLANE
0.0118 (0.30)
0.0040 (0.10)
0.0192 (0.49)
0.0138 (0.35)
0.1043 (2.65)
0.0926 (2.35)
0.0500
(1.27)
BSC
0.0125 (0.32)
0.0091 (0.23)
0.0500 (1.27)
0.0157 (0.40)
8
0
0.0291 (0.74)
0.0098 (0.25)
x 45