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Электронный компонент: 5962-9312901M2A

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CONNECTION DIAGRAMS
8-Lead Plastic Mini-DIP (N),
Cerdip (Q) and SOIC (R) Packages
TOP VIEW
(Not to Scale)
8
7
6
5
1
2
3
4
OFFSET NULL
C
COMP
AD829
IN
+IN
V
S
+V
S
OUTPUT
OFFSET NULL
20-Lead LCC Pinout
20 19
1
2
3
18
14
15
16
17
4
5
6
7
8
9 10 11 12 13
TOP VIEW
(Not to Scale)
AD829
NC
+V
NC
OUTPUT
NC
NC = NO CONNECT
NC
NC
NC
NC
V
NC
C
COMP
NC
OFFSET NULL
OFFSET NULL
NC
IN
NC
+IN
NC
REV. E
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
a
High-Speed, Low-Noise
Video Op Amp
AD829
FEATURES
High Speed
120 MHz Bandwidth, Gain = 1
230 V/ s Slew Rate
90 ns Settling Time to 0.1%
Ideal for Video Applications
0.02% Differential Gain
0.04 Differential Phase
Low Noise
1.7 nV/
Hz Input Voltage Noise
1.5 pA/
Hz Input Current Noise
Excellent DC Precision
1 mV max Input Offset Voltage (Over Temp)
0.3 V/ C Input Offset Drift
Flexible Operation
Specified for 5 V to 15 V Operation
3 V Output Swing into a 150 Load
External Compensation for Gains 1 to 20
5 mA Supply Current
Available in Tape and Reel in Accordance with
EIA-481A Standard
PRODUCT DESCRIPTION
The AD829 is a low noise (1.7 nV/
Hz), high speed op amp
with custom compensation that provides the user with gains
from
1 to 20 while maintaining a bandwidth greater than
50 MHz. The AD829's 0.04
differential phase and 0.02%
differential gain performance at 3.58 MHz and 4.43 MHz,
driving reverse-terminated 50
or 75 cables, makes it ideally
suited for professional video applications. The AD829 achieves
its 230 V/
s uncompensated slew rate and 750 MHz gain band-
width product while requiring only 5 mA of current from the
power supplies.
The AD829's external compensation pin gives it exceptional
versatility. For example, compensation can be selected to opti-
mize the bandwidth for a given load and power supply voltage.
As a gain-of-two line driver, the 3 dB bandwidth can be in-
creased to 95 MHz at the expense of 1 dB of peaking. In addi-
tion, the AD829's output can also be clamped at its external
compensation pin.
The AD829 has excellent dc performance. It offers a minimum
open-loop gain of 30 V/mV into loads as low as 500
, low
input voltage noise of 1.7 nV/
Hz, and a low input offset volt-
age of 1 mV maximum. Common-mode rejection and power
supply rejection ratios are both 120 dB.
The AD829 is also useful in multichannel, high speed data
conversion where its fast (90 ns to 0.1%) settling time is of
importance. In such applications, the AD829 serves as an input
buffer for 8-to-10-bit A/D converters and as an output I/V con-
verter for high speed D/A converters.
The AD829 provides many of the same advantages that a trans-
impedance amplifier offers, while operating as a traditional
voltage feedback amplifier. A bandwidth greater than 50 MHz
can be maintained for a range of gains by changing the external
compensation capacitor. The AD829 and the transimpedance
amplifier are both unity gain stable and provide similar voltage
noise performance (1.7 nV/
Hz). However, the current noise of
the AD829 (1.5 pA/
Hz) is less than 10% of the noise of trans-
impedance amps. Furthermore, the inputs of the AD829 are
symmetrical.
PRODUCT HIGHLIGHTS
1. Input voltage noise of 2 nV/
Hz, current noise of 1.5 pA/
Hz and 50 MHz bandwidth, for gains of 1 to 20, make the
AD829 an ideal preamp.
2. Differential phase error of 0.04
and a 0.02% differential
gain error, at the 3.58 MHz NTSC and 4.43 MHz PAL and
SECAM color subcarrier frequencies, make it an outstanding
video performer for driving reverse-terminated 50
and
75
cables to 1 V (at their terminated end).
3. The AD829 can drive heavy capacitive loads.
4. Performance is fully specified for operation from
5 V to
15 V supplies.
5. Available in plastic, cerdip, and small outline packages.
Chips and MIL-STD-883B parts are also available.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700
World Wide Web Site: http://www.analog.com
Fax: 781/326-8703
Analog Devices, Inc., 2000
REV. E
2
AD829SPECIFICATIONS
AD829J/AR
AD829AQ/S
Model
Conditions
V
S
Min
Typ
Max
Min
Typ
Max
Units
INPUT OFFSET VOLTAGE
5 V, 15 V
0.2
1
0.1
0.5
mV
T
MIN
to T
MAX
1
0.5
mV
Offset Voltage Drift
5 V, 15 V
0.3
0.3
V/C
INPUT BIAS CURRENT
5 V, 15 V
3.3
7
3.3
7
A
T
MIN
to T
MAX
8.2/9.5
9.5
A
INPUT OFFSET CURRENT
5 V, 15 V
50
500
50
500
nA
T
MIN
to T
MAX
500
500
nA
Offset Current Drift
5 V, 15 V
0.5
0.5
nA/
C
OPEN-LOOP GAIN
V
O
=
2.5 V
5 V
R
LOAD
= 500
30
65
30
65
V/mV
T
MIN
to T
MAX
20
20
V/mV
R
LOAD
= 150
40
40
V/mV
V
OUT
=
10 V
15 V
R
LOAD
= 1 k
50
100
50
100
V/mV
T
MIN
to T
MAX
20
20
V/mV
R
LOAD
= 500
85
85
V/mV
DYNAMIC PERFORMANCE
Gain Bandwidth Product
5 V
600
600
MHz
15 V
750
750
MHz
Full Power Bandwidth
1, 2
V
O
= 2 V p-p
R
LOAD
= 500
5 V
25
25
MHz
V
O
= 20 V p-p
R
LOAD
= 1 k
15 V
3.6
3.6
MHz
Slew Rate
2
R
LOAD
= 500
5 V
150
150
V/
s
R
LOAD
= 1 k
15 V
230
230
V/
s
Settling Time to 0.1%
A
V
= 19
2.5 V to +2.5 V
5 V
65
65
ns
10 V Step
15 V
90
90
ns
Phase Margin
2
C
LOAD
= 10 pF
15 V
R
LOAD
= 1 k
60
60
Degrees
DIFFERENTIAL GAIN ERROR
3
R
LOAD
= 100
15 V
C
COMP
= 30 pF
0.02
0.02
%
DIFFERENTIAL PHASE ERROR
3
R
LOAD
= 100
15 V
C
COMP
= 30 pF
0.04
0.04
Degrees
COMMON-MODE REJECTION
V
CM
=
2.5 V
5 V
100
120
100
120
dB
V
CM
=
12 V
15 V
100
120
100
120
dB
T
MIN
to T
MAX
96
96
dB
POWER SUPPLY REJECTION
V
S
=
4.5 V to 18 V
98
120
98
120
dB
T
MIN
to T
MAX
94
94
dB
INPUT VOLTAGE NOISE
f = 1 kHz
15 V
1.7
2
1.7
2
nV/
Hz
INPUT CURRENT NOISE
f = 1 kHz
15 V
1.5
1.5
pA/
Hz
INPUT COMMON-MODE
VOLTAGE RANGE
5 V
+4.3
+4.3
V
3.8
3.8
V
15 V
+14.3
+14.3
V
13.8
13.8
V
OUTPUT VOLTAGE SWING
R
LOAD
= 500
5 V
3.0
3.6
3.0
3.6
V
R
LOAD
= 150
5 V
2.5
3.0
2.5
3.0
V
R
LOAD
=
50
5 V
1.4
1.4
V
R
LOAD
= 1 k
15 V
12
13.3
12
13.3
V
R
LOAD
= 500
15 V
10
12.2
10
12.2
V
Short Circuit Current
5 V, 15 V
32
32
mA
INPUT CHARACTERISTICS
Input Resistance (Differential)
13
13
k
Input Capacitance (Differential)
4
5
5
pF
Input Capacitance (Common Mode)
1.5
1.5
pF
CLOSED-LOOP OUTPUT
RESISTANCE
A
V
= +1, f = 1 kHz
2
2
m
(@ T
A
= +25 C and V
S
= 15 V dc, unless otherwise noted)
AD829J/AR
AD829AQ/S
Model
Conditions
V
S
Min
Typ
Max
Min
Typ
Max
Units
POWER SUPPLY
Operating Range
4.5
18
4.5
18
V
Quiescent Current
5 V
5
6.5
5
6.5
mA
T
MIN
to T
MAX
8.0
8.2/8.7
mA
15 V
5.3
6.8
5.3
6.8
mA
T
MIN
to T
MAX
8.3/8.5
8.5/9.0
mA
TRANSISTOR COUNT
Number of Transistors
46
46
NOTES
1
Full Power Bandwidth = Slew Rate/2
V
PEAK
.
2
Tested at Gain = +20, C
COMP
= 0 pF.
3
3.58 MHz (NTSC) and 4.43 MHz (PAL & SECAM).
4
Differential input capacitance consists of 1.5 pF package capacitance plus 3.5 pF from the input differential pair.
Specifications subject to change without notice.
AD829
REV. E
3
ABSOLUTE MAXIMUM RATINGS
1
Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
18 V
Internal Power Dissipations
2
Plastic (N) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts
Small Outline (R) . . . . . . . . . . . . . . . . . . . . . . . . . 0.9 Watts
Cerdip (Q) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.3 Watts
LCC (E) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0.8 Watts
Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
V
S
Differential Input Voltage
3
. . . . . . . . . . . . . . . . . . . .
6 Volts
Output Short Circuit Duration . . . . . . . . . . . . . . . . Indefinite
Storage Temperature Range (Q, E) . . . . . . . 65
C to +150C
Storage Temperature Range (N, R) . . . . . . . 65
C to +125C
Operating Temperature Range
AD829J . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0
C to +70C
AD829A . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
C to +85C
AD829S . . . . . . . . . . . . . . . . . . . . . . . . . . 55
C to +125C
Lead Temperature Range (Soldering 60 sec) . . . . . . . . +300
C
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only and functional operation of
the device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Maximum internal power dissipation is specified so that T
J
does not exceed
+175
C at an ambient temperature of +25C.
Thermal characteristics:
8-lead plastic package:
JA
= 100
C/watt (derate at 8.7 mW/C)
8-lead cerdip package:
JA
= 110
C/watt (derate at 8.7 mW/C)
20-lead LCC package:
JA
= 150
C/watt
8-lead small outline package:
JA
= 155
C/watt (derate at 6 mW/C).
3
If the differential voltage exceeds 6 volts, external series protection resistors should
be added to limit the input current.
ORDERING GUIDE
Model
Temperature Range
Package Description
Package Option*
AD829JN
0
C to +70C
8-Lead Plastic Mini-DIP
N-8
AD829AR
40
C to +85C
8-Lead Plastic SOIC
SO-8
AD829JR
0
C to +70C
8-Lead Plastic SOIC
SO-8
AD829AR-REEL7
40
C to +85C
Tape and Reel 7"
AD829AR-REEL
40
C to +85C
Tape and Reel 13"
AD829JR-REEL7
0
C to +70C
Tape and Reel 7"
AD829JR-REEL
0
C to +70C
Tape and Reel 13"
AD829AQ
40
C to +85C
8-Lead Cerdip
Q-8
AD829SQ
55
C to +125C
8-Lead Cerdip
Q-8
AD829SQ/883B
55
C to +125C
8-Lead Cerdip
Q-8
5962-9312901MPA
55
C to +125C
8-Lead Cerdip
Q-8
AD829SE/883B
55
C to +125C
20-Lead LCC
E-20A
5962-9312901M2A
55
C to +125C
20-Lead LCC
E-20A
AD829JCHIPS
0
C to +70C
Die
AD829SCHIPS
55
C to +125C
Die
*E = Leadless Chip Carrier (Ceramic); N = Plastic DIP; Q = Cerdip; SO = Small Outline IC (SOIC).
ESD SUSCEPTIBILITY
ESD (electrostatic discharge) sensitive device. Electrostatic
charges as high as 4000 volts, which readily accumulate on the
human body and on test equipment, can discharge without
detection. Although the AD829 features proprietary ESD pro-
tection circuitry, permanent damage may still occur on these
devices if they are subjected to high energy electrostatic dis-
charges. Therefore, proper ESD precautions are recommended
to avoid any performance degradation or loss of functionality.
METALIZATION PHOTO
Contact factory for latest dimensions.
Dimensions shown in inches and (mm).
AD829Typical Performance Characteristics
REV. E
4
SUPPLY VOLTAGE Volts
INPUT COMMON-MODE RANGE
Volts
20
15
0
10
5
0
20
5
10
15
+V
OUT
V
OUT
Figure 1. Input Common-Mode
Range vs. Supply Voltage
QUIESCENT CURRENT
mA
SUPPLY VOLTAGE Volts
6.0
5.5
4.0
5.0
4.5
0
20
5
10
15
Figure 4. Quiescent Current vs.
Supply Voltage
7
6
5
4
3
QUIESCENT CURRENT
mA
60
20
0
20
40
60
80 100
140
40
120
TEMPERATURE C
V
S
= 5V
V
S
= 15V
Figure 7. Quiescent Current vs.
Temperature
MAGNITUDE OF THE OUTPUT
VOLTAGE
Volts
SUPPLY VOLTAGE Volts
20
15
0
10
5
0
20
5
10
15
+V
OUT
V
OUT
R
LOAD
= 1k
Figure 2. Output Voltage Swing
vs. Supply Voltage
5
4
3
2
60
20
0
20
40
60
80 100
140
40
120
INPUT BIAS CURRENT
A
TEMPERATURE C
V
S
= 5V, 15V
Figure 5. Input Bias Current vs.
Temperature
40
35
30
25
20
15
SHORT CIRCUIT CURRENT LIMIT
mA
60
20
0
20
40
60
80 100
140
40
120
AMBIENT TEMPERATURE C
V
S
= 5V
NEGATIVE
CURRENT LIMIT
POSITIVE
CURRENT LIMIT
Figure 8. Short Circuit Current
Limit vs. Temperature
OUTPUT VOLTAGE SWING
Volts p
p
30
25
20
15
10
5
0
10
100
1k
10k
LOAD RESISTANCE
5 VOLT
SUPPLIES
15 VOLT
SUPPLIES
Figure 3. Output Voltage Swing
vs. Resistive Load
100
10
1
0.1
0.01
1k
10k
100k
1M
10M
100M
FREQUENCY Hz
0.001
CLOSED - LOOP OUTPUT IMPEDANCE
A
V
= +1
C
COMP
= 68pF
A
V
= +20
C
COMP
= 0pF
Figure 6. Closed-Loop Output
Impedance vs. Frequency
65
60
55
50
45
3 dB BANDWIDTH
MHz
60
20
0
20
40
60
80 100
140
40
120
TEMPERATURE C
V
S
= 15V
A
V
= +20
C
COMP
= 0pF
Figure 9. 3 dB Bandwidth vs.
Temperature
AD829
REV. E
5
120
100
80
60
40
20
0
100
1k
10k
100k
1M
10M
100M
+100
+80
+60
+40
20
0
PHASE
Degrees
OPEN-LOOP GAIN
dB
FREQUENCY Hz
+20
GAIN
15V
Supplies
1k Load
GAIN
5V
Supplies
500 Load
PHASE
C
COMP
= 0pF
Figure 10. Open-Loop Gain & Phase
Margin vs. Frequency
120
100
80
60
40
CMRR
dB
1k
10k
100k
1M
10M
100M
FREQUENCY Hz
20
C
COMP
= 0pF
Figure 13. Common-Mode Rejection
Ratio vs. Frequency
100
300
1k
3k
10k
30k
100k
THD
dB
110
FREQUENCY Hz
105
100
95
90
85
80
75
70
R
L
= 500
R
L
= 2k
V
IN
= 3V RMS
A
V
= 1
C
COMP
= 30pF
C
LOAD
= 100pF
Figure 16. Total Harmonic Dis-
tortion (THD) vs. Frequency
105
100
95
90
85
80
75
OPEN-LOOP GAIN
dB
10
100
1k
10k
LOAD RESISTANCE
V
S
= 5V
V
S
= 15V
Figure 11. Open-Loop Gain vs.
Resistive Load
30
25
20
15
10
5
0
OUTPUT VOLTAGE
Volts p
p
10
100
INPUT FREQUENCY MHz
1
V
S
= 15V
R
L
= 1k
A
V
= +20
C
COMP
= 0pF
V
S
= 5V
R
L
= 500
A
V
= +20
C
COMP
= 0pF
Figure 14. Large Signal Frequency
Response
2nd HARMONIC
20
30
40
50
60
70
V
IN
= 2.24V RMS
A
V
= 1
R
L
= 250
C
LOAD
= 0
C
COMP
= 30pF
THD
dB
0
500k
1M
1.5M
2M
FREQUENCY Hz
3rd HARMONIC
Figure 17. 2nd & 3rd Harmonic
Distortion vs. Frequency
120
100
80
60
40
PSRR
dB
1k
10k
100k
1M
10M
100M
FREQUENCY Hz
20
SUPPLY
+
SUPPLY
C
COMP
= 0pF
Figure 12. Power Supply Rejection
Ratio (PSRR) vs. Frequency
10
8
6
4
2
0
0
20
40
60
80
100 120
160
1%
1%
140
2
4
6
8
0.1%
ERROR
A
V
= 19
C
COMP
= 0pF
10
OUTPUT SWING FROM 0 TO
V
SETTLING TIME ns
0.1%
Figure 15. Output Swing & Error vs.
Settling Time
5
4
3
2
1
0
INPUT VOLTAGE NOISE
nV
/
Hz
10
100
1k
10k
100k
FREQUENCY Hz
1M
10M
Figure 18. Input Voltage Noise
Spectral Density
AD829Typical Performance Characteristics
REV. E
6
400
350
300
250
200
150
100
A
V
= +20
SLEW RATE 10 90%
SLEW RATE
Volts /
s
60
20
0
20
40
60
80 100
140
40
120
TEMPERATURE C
V
S
= 5V
V
S
= 15V
RISE
FALL
RISE
FALL
Figure 19. Slew Rate vs. Temperature
0.05
0.04
5
10
15
0.03
SUPPLY VOLTAGE Volts
0.02
0.01
0.03
DIFFERENTIAL PHASE
Degrees
DIFFERENTIAL GAIN
Percent
DIFF PHASE
0.043
DIFF GAIN
Figure 20. Differential Gain & Phase
vs. Supply
+V
S
0.1 F
C
COMP
(EXTERNAL)
V
S
0.1 F
OFFSET
NULL
ADJUST
20k
AD829
5pF
300
300
50
HP8130A
5ns RISE TIME
50
CABLE
TEKTRONIX
TYPE 7A24
PREAMP
+15V
0.1 F
C
COMP
15pF
15V
0.1 F
AD829
50
CABLE
50
50
Figure 22a. Follower Connection. Gain = +2
Figure 22c. Gain-of-2 Follower
Small Signal Pulse Response
Figure 22b. Gain-of-2 Follower
Large Signal Pulse Response
Figure 21. Offset Null and External
Shunt Compensation Connections
AD829
REV. E
7
50
HP8130A
5ns RISE TIME
50
CABLE
TEKTRONIX
TYPE 7A24
PREAMP
+15V
0.1 F
C
COMP
15pF
15V
0.1 F
AD829
50
CABLE
50
50
300
300
5pF
Figure 24a. Unity Gain Inverter Connection
1pF
2k
105
5
HP8130A
5ns RISE TIME
50
CABLE
TEKTRONIX
TYPE 7A24
PREAMP
45
100
FET PROBE
+15V
0.1 F
C
COMP
= 0pF
15V
0.1 F
AD829
Figure 23a. Follower Connection. Gain = +20
Figure 23b. Gain-of-20 Follower
Large Signal Pulse Response
Figure 23c. Gain-of-20 Follower
Small Signal Pulse Response
Figure 24c. Unity Gain Inverter
Small Signal Pulse Response
Figure 24b. Unity Gain Inverter
Large Signal Pulse Response
AD829
REV. E
8
THEORY OF OPERATION
The AD829 is fabricated on Analog Devices' proprietary comple-
mentary bipolar (CB) process which provides PNP and NPN
transistors with similar f
T
s of 600 MHz. As shown in Figure 25,
the AD829 input stage consists of an NPN differential pair in
which each transistor operates at 600
A collector current. This
gives the input devices a high transconductance and hence gives
the AD829 a low noise figure of 2 nV/
Hz @ 1 kHz.
The input stage drives a folded cascode which consists of a fast
pair of PNP transistors. These PNPs then drive a current mirror
which provides a differential-input to single-ended-output con-
version. The high speed PNPs are also used in the current-
amplifying output stage which provides high current gain of
40,000. Even under conditions of heavy loading, the high f
T
s
of the NPN & PNPs, produced using the CB process, permit
cascading two stages of emitter followers while still maintaining
60
of phase margin at closed-loop bandwidths greater than
50 MHz.
Two stages of complementary emitter followers also effectively
buffer the high impedance compensation node (at the C
COMP
pin) from the output so that the AD829 can maintain a high dc
open-loop gain, even into low load impedances: 92 dB into a
150
load, 100 dB into a 1 k load. Laser trimming and
PTAT biasing assure low offset voltage and low offset voltage
drift enabling the user to eliminate ac coupling in many
applications.
For added flexibility, the AD829 provides access to the internal
frequency compensation node. This allows the user to customize
frequency response characteristics for a particular application.
Unity gain stability requires a compensation capacitance of
68 pF (Pin 5 to ground) which will yield a small signal band-
width of 66 MHz and slew rate of 16 V/
s. The slew rate and
gain bandwidth product will vary inversely with compensation
capacitance. Table I and the graph of Figure 28 show the opti-
mum compensation capacitance and the resulting slew rate for a
desired noise gain. For gains between 1 and 20, C
COMP
can be
chosen to keep the small signal bandwidth relatively constant.
The minimum gain which will still provide stability also de-
pends on the value of external compensation capacitance.
An RC network in the output stage (Figure 25) completely
removes the effect of capacitive loading when the amplifier is
compensated for closed-loop gains of 10 or higher. At low fre-
quencies, and with low capacitive loads, the gain from the com-
pensation node to the output is very close to unity. In this case,
C is bootstrapped and does not contribute to the compensation
capacitance of the device. As the capacitive load is increased, a
pole is formed with the output impedance of the output stage
this reduces the gain, and subsequently, C is incompletely boot-
strapped. Therefore, some fraction of C contributes to the
compensation capacitance, and the unity gain bandwidth falls.
As the load capacitance is further increased, the bandwidth
continues to fall, and the amplifier remains stable.
Externally Compensating the AD829
The AD829 is stable with no external compensation for noise
gains greater than 20. For lower gains, there are two methods of
frequency compensating the amplifier to achieve closed-loop
stability; these are the shunt and current feedback compensation
methods.
IN
IN
+
1.2mA
OFFSET NULL
15
C
12.5pF
R
500
OUTPUT
+V
S
V
S
C
COMP
15
Figure 25. AD829 Simplified Schematic
Shunt Compensation
Figures 26 and 27 show that the first method, shunt compensa-
tion, has an external compensation capacitor, C
COMP
, connected
between the compensation pin and ground. This external
capacitor is tied in parallel with approximately 3 pF of inter-
nal capacitance at the compensation node. In addition, a
small capacitance, C
LEAD
, in parallel with resistor R2, compen-
sates for the capacitance at the amplifier's inverting input.
50
50
COAX
CABLE
V
S
0.1 F
C
COMP
0.1 F
AD829
C
LEAD
R2
V
OUT
R1
+V
S
V
IN
1k
Figure 26. Inverting Amplifier Connection Using External
Shunt Compensation
50
50
CABLE
V
S
0.1 F
C
COMP
0.1 F
AD829
V
OUT
+V
S
V
IN
1k
R2
C
LEAD
R1
Figure 27. Noninverting Amplifier Connection Using
External Shunt Compensation
AD829
REV. E
9
Table I. Component Selection for Shunt Compensation
Slew
3 dB
Follower
Inverter
R1
R2
C
L
C
COMP
Rate
Small Signal
Gain
Gain
pF
pF
V/ s
Bandwidth MHz
1
Open
100
0
68
16
66
2
1
1k
1k
5
25
38
71
5
4
511
2.0k
1
7
90
76
10
9
226
2.05k
0
3
130
65
20
19
105
2k
0
0
230
55
25
24
105
2.49
0
0
230
39
100
99
20
2k
0
0
230
7.5
Table I gives recommended C
COMP
and C
LEAD
values along with
the corresponding slew rates and bandwidth. The capacitor
values given were selected to provide a small signal frequency
response with less than 1 dB of peaking and less than 10% over-
shoot. For this table, supply voltages of
15 volts should be
used. Figure 28 is a graphical extension of the table which
shows the slew rate/gain trade-off for lower closed-loop gains,
when using the shunt compensation scheme.
NOISE GAIN
100
1
1
100
10
C
COMP

pF
10
10
100
1k
SLEW RATE = V/
s
SLEW RATE
V
S
= 15V
C
COMP
Figure 28. Value of C
COMP
& Slew Rate vs. Noise Gain
Current Feedback Compensation
Bipolar nondegenerated amplifiers which are single pole and
internally compensated have their bandwidths defined as:
f
T
=
1
2
r
e
C
COMP
=
I
2
kT
q
C
COMP
where:
f
T
is the unity gain bandwidth of the amplifier
I is the collector current of the input transistor
C
COMP
is the compensation capacitance
r
e
is the inverse of the transconductance of the input transistors
kT/q is approximately equal to 26 mV @ 27
C.
Since both f
T
and slew rate are functions of the same variables,
the dynamic behavior of an amplifier is limited. Since:
Slew Rate
=
2I
C
COMP
then:
Slew Rate
f
T
= 4
kT
q
This shows that the slew rate will be only 0.314 V/
s for every
MHz of bandwidth. The only way to increase slew rate is to
increase the f
T
and that is difficult, due to process limitations.
Unfortunately, an amplifier with a bandwidth of 10 MHz can
only slew at 3.1 V/
s, which is barely enough to provide a full
power bandwidth of 50 kHz.
The AD829 is especially suited to a new form of compensation
which allows for the enhancement of both the full power band-
width and slew rate of the amplifier. The voltage gain from the
inverting input pin to the compensation pin is large; therefore, if
a capacitance is inserted between these pins, the amplifier's
bandwidth becomes a function of its feedback resistor and this
capacitance. The slew rate of the amplifier is now a function of
its internal bias (2I) and this compensation capacitance.
Since the closed-loop bandwidth is a function of R
F
and C
COMP
(Figure 29), it is independent of the amplifier closed-loop gain,
as shown in Figure 31. To preserve stability, the time constant
of R
F
and C
COMP
needs to provide a bandwidth of less than
65 MHz. For example, with C
COMP
= 15 pF and R
F
= 1 k
, the
small signal bandwidth of the AD829 is 10 MHz, while Figure
30 shows that the slew rate is in excess of 60 V/
s. As can be
seen in Figure 31, the closed-loop bandwidth is constant for
gains of 1 to 4, a property of current feedback amplifiers.
FOR C
1
OF C
COMP
C
COMP
SHOULD NEVER EXCEED
15pF FOR THIS CONNECTION
0pF
15pF
<7pF
7pF
50
50
COAX
CABLE
V
S
0.1 F
0.1 F
AD829
R
F
V
OUT
R1
+V
S
V
IN
R
L
1k
IN4148
C
COMP
C
1
*
*RECOMMENDED VALUE
Figure 29. Inverting Amplifier Connection Using Current
Feedback Compensation
AD829
REV. E
10
Figure 30. Large Signal Pulse Response of Inverting
Amplifier Using Current Feedback Compensation.
C
COMP
= 15 pF, C1 = 15 pF, R
F
= 1 k
, R1 = 1 k
FREQUENCY Hz
15
100k
100M
CLOSED-LOOP GAIN
dB
1M
10M
12
9
6
3
0
3
6
9
12
15
3dB @ 8.2MHz
3dB @ 10.2MHz
3dB @ 9.6MHz
GAIN = 2
V
IN
= 30dBM
V
S
= 15V
R
L
= 1k
R
F
= 1k
C
COMP
= 15pF
C
1
= 15pF
GAIN = 4
GAIN = 1
Figure 31. Closed-Loop Gain vs. Frequency for the Circuit
of Figure 29
Figure 32 is an oscilloscope photo of the pulse response of a
unity gain inverter which has been configured to provide a small
signal bandwidth of 53 MHz and a subsequent slew rate of
180 V/
s; resistor R
F
= 3 k
, capacitor C
COMP
= 1 pF. Figure 33
shows the excellent pulse response as a unity gain inverter, this
time using component values of: R
F
= 1 k
and C
COMP
= 4 pF.
Figures 34 and 35 show the closed-loop frequency response of
the AD829 for different closed-loop gains and for different
supply voltages.
If a noninverting amplifier configuration using current feedback
compensation is desired, the circuit of Figure 36 is recom-
mended. This circuit doubles the slew rate compared to the
shunt compensated noninverting amplifier of Figure 27 at the
expense of gain flatness. Nonetheless, this circuit delivers 95 MHz
bandwidth with
1 dB flatness into a back terminated cable,
with a differential gain error of only 0.01%, and a differential
phase error of only 0.015
at 4.43 MHz.
Figure 32. Large Signal Pulse Response of the Inverting
Amplifier Using Current Feedback Compensation.
C
COMP
= 1 pF, R
F
= 3 k
, R1 = 3 k
Figure 33. Small Signal Pulse Response of Inverting
Amplifier Using Current Feedback Compensation.
C
COMP
= 4 pF, R
F
= 1 k
, R1 = 1 k
FREQUENCY Hz
15
100M
CLOSED-LOOP GAIN
dB
1M
10M
12
9
6
3
0
3
6
9
12
15
GAIN = 2
V
S
= 15V
R
L
= 1k
R
F
= 1k
V
IN
= 30dBM
GAIN = 4
GAIN = 1
C
COMP
= 2pF
C
COMP
= 3pF
C
COMP
= 4pF
Figure 34. Closed-Loop Frequency Response for the
Inverting Amplifier Using Current Feedback Compensation
AD829
REV. E
11
+15V
15V
COMM
10 F
EACH
AMPLIFIER
10 F
0.1 F
0.1 F
+V
S
1 F
1 F
0.1 F
0.1 F
V
S
PIN 4
PIN 7
INPUT
FREQUENCY
CMRR
100 Hz
1 MHz
10 MHz
64.6dB
44.7dB
23.9dB
AD829
+V
IN
28pF
SETTLING TIME
AC CMR ADJUST
3pF
AD829
3pF
1pF
1pF
2k
2k
R
G
210
AD848
(G = 5)
A1
A2
A3
+V
IN
200
200
970
50
DC CMR
ADJUST
1k
3pF
2k
(G = 20)
(G = 20)
CIRCUIT GAIN =
(
4000
R
G
+ 1
(
5
Figure 38. A High Gain, Video Bandwidth Three Op Amp In Amp Circuit
FREQUENCY Hz
47
100M
OUTPUT LEVEL
dB
1M
10M
44
41
38
35
32
29
26
23
20
17
V
IN
= 20dBM
R
L
= 1k
R
F
= 1k
GAIN = 1
C
COMP
= 4pF
15V
5V
Figure 35. Closed-Loop Frequency Response vs. Supply
for the Inverting Amplifier Using Current Feedback
Compensation
A Low Error Video Line Driver
The buffer circuit shown in Figure 37 will drive a back-termi-
nated 75
video line to standard video levels (1 V p-p) with
0.1 dB gain flatness to 30 MHz with only 0.04
and 0.02%
differential phase and gain at the 4.43 MHz PAL color
subcarrier frequency. This level of performance, which meets
the requirements for high definition video displays and test
equipment, is achieved using only 5 mA quiescent current.
A High Gain, Video Bandwidth Three Op Amp In Amp
Figure 38 shows a three op amp instrumentation amplifier cir-
cuit which provides a gain of 100 at video bandwidths. At a
circuit gain of 100 the small signal bandwidth equals 18 MHz
into an FET probe. Small signal bandwidth equals 6.6 MHz
with a 50
load. 0.1% settling time is 300 ns.
50
50
COAX
CABLE
0.1 F
3pF
C
COMP
0.1 F
AD829
V
OUT
+15V
V
IN
15V
50
50
COAX
CABLE
50
2k
2k
Figure 36. Noninverting Amplifier Connection Using
Current Feedback Compensation
0.1 F
30pF
0.1 F
AD829
V
OUT
+15V
V
IN
15V
75
75
COAX
CABLE
75
300
OPTIONAL
2 7pF
FLATNESS
TRIM
C
COMP
300
75
Figure 37. A Video Line Driver with a Flatness over
Frequency Adjustment
The input amplifiers operate at a gain of 20, while the output
op amp runs at a gain of 5. In this circuit the main bandwidth
limitation is the gain/ bandwidth product of the output ampli-
fier. Extra care needs to be taken while breadboarding this cir-
cuit, since even a couple of extra picofarads of stray capacitance
at the compensation pins of A1 and A2 will degrade circuit
bandwidth.
AD829
REV. E
12
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
Cerdip (Q) Package
0.320 (8.13)
0.290 (7.37)
0.015 (0.38)
0.008 (0.20)
15
0
0.005 (0.13) MIN
0.055 (1.40) MAX
1
PIN 1
4
5
8
0.310 (7.87)
0.220 (5.59)
0.405 (10.29) MAX
0.200
(5.08)
MAX
SEATING
PLANE
0.023 (0.58)
0.014 (0.36)
0.070 (1.78)
0.030 (0.76)
0.060 (1.52)
0.015 (0.38)
0.150
(3.81)
MIN
0.200 (5.08)
0.125 (3.18)
0.100
(2.54)
BSC
C1443c05/00 (rev. E) 00880
PRINTED IN U.S.A.
Plastic Mini-DIP (N) Package
0.0110.003
(0.280.08)
0.30 (7.62)
REF
15
0
PIN 1
4
5
8
1
0.25
(6.35)
0.31
(7.87)
0.10
(2.54)
BSC
SEATING
PLANE
0.0350.01
(0.890.25)
0.180.03
(4.570.76)
0.033
(0.84)
NOM
0.0180.003
(0.460.08)
0.125
(3.18)
MIN
0.1650.01
(4.190.25)
0.39 (9.91) MAX
8-Lead SOIC (R) Package
8
5
4
1
0.1968 (5.00)
0.1890 (4.80)
0.2440 (6.20)
0.2284 (5.80)
PIN 1
0.1574 (4.00)
0.1497 (3.80)
0.0500 (1.27)
BSC
0.0688 (1.75)
0.0532 (1.35)
SEATING
PLANE
0.0098 (0.25)
0.0040 (0.10)
0.0192 (0.49)
0.0138 (0.35)
0.0098 (0.25)
0.0075 (0.19)
0.0500 (1.27)
0.0160 (0.41)
8
0
0.0196 (0.50)
0.0099 (0.25)
45
All brand or product names mentioned are trademarks or registered trademarks of their respective holders.
20-Lead LCC (E-20A) Package
1
20
4
9
8
13
19
BOTTOM
VIEW
14
3
18
0.028 (0.71)
0.022 (0.56)
45 TYP
0.015 (0.38)
MIN
0.055 (1.40)
0.045 (1.14)
0.050 (1.27)
BSC
0.075 (1.91)
REF
0.011 (0.28)
0.007 (0.18)
R TYP
0.095 (2.41)
0.075 (1.90)
0.100 (2.54) BSC
0.200 (5.08)
BSC
0.150 (3.81)
BSC
0.075
(1.91)
REF
0.358 (9.09)
0.342 (8.69)
SQ
0.358
(9.09)
MAX
SQ
0.100 (2.54)
0.064 (1.63)
0.088 (2.24)
0.054 (1.37)