PIN CONNECTIONS

8-Lead Narrow-Body SO

8-Lead Epoxy DIP

(S Suffix)

(P Suffix)

OUT A

IN A

+IN A

V+

OUT B

IN B

+IN B

OP295

OUT A

IN A

+IN A

V+

OUT B

IN B

+IN B

OP295

14-Lead Epoxy DIP

16-Lead SO (300 Mil)

(P Suffix)

(S Suffix)

OUT A

IN A

+IN A

OUT B

IN D

+IN D

OUT D

IN B

+IN B

V+

OUT C

IN C

+IN C

OP495

OUT A

IN A

+IN A

OUT D

IN D

+IN D

OUT B

IN B

+IN B

V+

OUT C

IN C

+IN C

OP495

NC = NO CONNECT

REV. B

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

Dual/Quad Rail-to-Rail
Operational Amplifiers

OP295/OP495

One Technology Way, P.O. Box 9106, Norwood. MA 02062-9106, U.S.A.
Tel: 617/329-4700

Fax: 617/326-8703

FEATURES
Rail-to-Rail Output Swing
Single-Supply Operation: +3 V to +36 V
Low Offset Voltage: 300 V
Gain Bandwidth Product: 75 kHz
High Open-Loop Gain: 1000 V/mV
Unity-Gain Stable
Low Supply Current/Per Amplifier: 150 A max

APPLICATIONS
Battery Operated Instrumentation
Servo Amplifiers
Actuator Drives
Sensor Conditioners
Power Supply Control

GENERAL DESCRIPTION

Rail-to-rail output swing combined with dc accuracy are the key
features of the OP495 quad and OP295 dual CBCMOS opera-
tional amplifiers. By using a bipolar front end, lower noise and
higher accuracy than that of CMOS designs has been achieved.
Both input and output ranges include the negative supply, pro-
viding the user "zero-in/zero-out" capability. For users of 3.3
volt systems such as lithium batteries, the OP295/OP495 is
specified for three volt operation.

Maximum offset voltage is specified at 300

V for +5 volt opera-

tion, and the open-loop gain is a minimum of 1000 V/mV. This
yields performance that can be used to implement high accuracy
systems, even in single supply designs.

The ability to swing rail-to-rail and supply +15 mA to the load
makes the OP295/OP495 an ideal driver for power transistors
and "H" bridges. This allows designs to achieve higher efficien-
cies and to transfer more power to the load than previously pos-
sible without the use of discrete components. For applications

that require driving inductive loads, such as transformers, in-
creases in efficiency are also possible. Stability while driving
capacitive loads is another benefit of this design over CMOS
rail-to-rail amplifiers. This is useful for driving coax cable or
large FET transistors. The OP295/OP495 is stable with loads in
excess of 300 pF.

The OP295 and OP495 are specified over the extended indus-
trial (40

C to +125

C) temperature range. OP295s are avail-

able in 8-pin plastic and ceramic DIP plus SO-8 surface mount
packages. OP495s are available in 14-pin plastic and SO-16
surface mount packages. Contact your local sales office for
MIL-STD-883 data sheet.

REV. B

OP295/OP495SPECIFICATIONS

ELECTRICAL CHARACTERISTICS

Parameter

Symbol

Conditions

Min

Typ

Max

Units

INPUT CHARACTERISTICS

Offset Voltage

300

+125

800

Input Bias Current

+125

Input Offset Current

+125

Input Voltage Range

+4.0

Common-Mode Rejection Ratio

CMRR

0 V

4.0 V, 40

+125

110

Large Signal Voltage Gain

= 10 k

, 0.005

OUT

4.0 V

1000

10,000

V/mV

= 10 k

, 40

+125

500

V/mV

Offset Voltage Drift

OUTPUT CHARACTERISTICS

Output Voltage Swing High

= 100 k

to GND

4.98

5.0

= 10 k

to GND

4.90

4.94

OUT

= 1 mA, 40

+125

4.7

Output Voltage Swing Low

= 100 k

to GND

0.7

= 10 k

to GND

0.7

OUT

= 1 mA, 40

+125

Output Current

OUT

POWER SUPPLY

Power Supply Rejection Ratio

PSRR

1.5 V

15 V

110

1.5 V

15 V,

+125

Supply Current Per Amplifier

OUT

= 2.5 V, R

, 40

+125

150

DYNAMIC PERFORMANCE

Skew Rate

= 10 k

0.03

Gain Bandwidth Product

GBP

kHz

Phase Margin

Degrees

NOISE PERFORMANCE

Voltage Noise

p-p

0.1 Hz to 10 Hz

1.5

V p-p

Voltage Noise Density

f = 1 kHz

nV/

Current Noise Density

f = 1 kHz

<0.1

pA/

Specifications subject to change without notice.

ELECTRICAL CHARACTERISTICS

Parameter

Symbol

Conditions

Min

Typ

Max

Units

INPUT CHARACTERISTICS

Offset Voltage

500

Input Bias Current

Input Offset Current

Input Voltage Range

+2.0

Common-Mode Rejection Ratio

CMRR

0 V

2.0 V, 40

+125

110

Large Voltage Gain

= 10 k

750

V/mV

Offset Voltage Drift

OUTPUT CHARACTERISTICS

Output Voltage Swing High

= 10 k

to GND

2.9

Output Voltage Swing Low

= 10 k

to GND

0.7

POWER SUPPLY

Power Supply Rejection Ratio

PSRR

1.5 V

15 V

110

1.5 V

15 V,

+125

Supply Current Per Amplifier

OUT

= 1.5 V, R

, 40

+125

150

DYNAMIC PERFORMANCE

Slew Rate

= 10 k

0.03

Gain Bandwidth Product

GBP

kHz

Phase Margin

Degrees

NOISE PERFORMANCE

Voltage Noise

p-p

0.1 Hz to 10 Hz

1.6

V p-p

Voltage Noise Density

f = 1 kHz

nV/

Current Noise Density

f = 1 kHz

<0.1

pA/

Specifications subject to change without notice.

(@ V

= +5.0 V, V

= +2.5 V, T

= +25 C unless otherwise noted)

(@ V

= +3.0 V, V

= +1.5 V, T

= +25 C unless otherwise noted)

ELECTRICAL CHARACTERISTICS

Parameter

Symbol

Conditions

Min

Typ

Max

Units

INPUT CHARACTERISTICS

Offset Voltage

300

+125

800

Input Bias Current

= 0 V

= 0 V, 40

+125

Input Offset Current

= 0 V

= 0 V, 40

+125

Input Voltage Range

+13.5

Common-Mode Rejection Ratio

CMRR

15.0 V

+13.5 V, 40

+125

110

Large Signal Voltage Gain

= 10 k

1000

4000

V/mV

Offset Voltage Drift

OUTPUT CHARACTERISTICS

Output Voltage Swing High

= 100 k

to GND

14.95

= 10 k

to GND

14.80

Output Voltage Swing Low

= 100 k

to GND

14.95

= 10 k

to GND

14.85

Output Current

OUT

POWER SUPPLY

Power Supply Rejection Ratio

PSRR

1.5 V to

15 V

110

1.5 V to

15 V, 40

+125

Supply Current

= 0 V, R

, V

18 V,

+125

175

Supply Voltage Range

+3 (

1.5)

+36 (

18)

DYNAMIC PERFORMANCE

Slew Rate

= 10 k

0.03

Gain Bandwidth Product

GBP

kHz

Phase Margin

Degrees

NOISE PERFORMANCE

Voltage Noise

n p-p

0.1 Hz to 10 Hz

1.25

V p-p

Voltage Noise Density

f =1 kHz

nV/

Current Noise Density

f = 1 kHz

<0.1

pA/

Specifications subject to change without notice.

WAFER TEST LIMITS

Parameter

Symbol

Conditions

Limit

Units

Offset Voltage

Vos

300

V max

Input Bias Current

nA max

Input Offset Current

nA max

Input Voltage Range

0 to +4

V min

Common-Mode Rejection Ratio

CMRR

0 V

4 V

dB min

Power Supply Rejection Ratio

PSRR

1.5 V

15 V

V/V

Large Signal Voltage Gain

= 10 k

1000

V/mV min

Output Voltage Swing High

= 10 k

4.9

V min

Supply Current Per Amplifier

OUT

= 2.5 V, R

150

A max

NOTES
Electrical tests and wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard
product dice. Consult factory to negotiate specifications based on dice lot qualifications through sample lot assembly and testing.

Guaranteed by CMRR test.

ORDERING GUIDE

OP295/OP495

REV. B

(@ V

15.0 V, T

= +25 C unless otherwise noted)

Temperature

Package

Model

Range

Description

Option

OP295GP

C to +125

8-Pin Plastic DIP

N-8

OP295GS

C to +125

8-Pin SOIC

SO-8

OP295GBC +25

DICE

Temperature

Package

Model

Range

Description

Option

OP495GP

C to +125

14-Pin Plastic DIP

N-14

OP495GS

C to +125

16-Pin SOL

R-16

OP495GBC +25

DICE

(@ V

= +5.0 V, V

= 2.5 V, T

= +25 C unless otherwise noted)

REV. B

OP295/OP495

ABSOLUTE MAXIMUM RATINGS

Supply Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

18 V

Input Voltage

. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .

18 V

Differential Input Voltage

. . . . . . . . . . . . . . . . . . . . . . . +36 V

Output Short-Circuit Duration . . . . . . . . . . . . . . . . . Indefinite
Storage Temperature Range

P, S Package . . . . . . . . . . . . . . . . . . . . . . . . 65

C to +150

Operating Temperature Range

OP295G, OP495G . . . . . . . . . . . . . . . . . . . 40

C to +125

Junction Temperature Range

P, S Package . . . . . . . . . . . . . . . . . . . . . . . . 65

C to +150

Lead Temperature Range (Soldering, 60 Sec) . . . . . . . +300

Package Type

Unit

8-Pin Plastic DIP (P)

103

C/W

8-Pin SOIC (S)

158

C/W

14-Pin Plastic DIP (P)

C/W

16-Pin SO (S)

C/W

NOTES

Absolute maximum ratings apply to both DICE and packaged parts, unless
otherwise noted.

For supply voltages less than

18 V, the absolute maximum input voltage is equal

to the supply voltage.

is specified for the worst case conditions, i.e.,

is specified for device in socket

for cerdip, P-DIP, and LCC packages;

is specified for device soldered in circuit

board for SOIC package.

DICE CHARACTERISTICS

OP295 Die Size 0.066

0.080 inch, 5,280 sq. mils.

Substrate (Die Backside) Is Connected to V+.
Transistor Count, 74.

OP495 Die Size 0.113

0.083 inch, 9,380 sq. mils.

Substrate (Die Backside) Is Connected to V+.
Transistor Count, 196.

Typical Characteristics

140

100

25

50

120

100

TEMPERATURE 

SUPPLY CURRENT PER AMPLIFIER 

= +5V

= +3V

= +36V

15.2

100

14.6

15.0

25

14.8

50

14.2

14.4

14.6

14.8

15.0

TEMPERATURE 

 OUTPUT SWING Volts

+ OUTPUT SWING Volts

15V

= 100k

= 2k

= 100k

= 10k

Supply Current Per Amplifier vs. Temperature

Output Voltage Swing vs. Temperature

REV. B

Typical CharacteristicsOP295/OP495

3.10

2.50

100

2.80

2.60

25

2.70

50

3.00

2.90

TEMPERATURE 

OUTPUT VOLTAGE SWING Volts

= 2k

= +3V

= 100k

= 10k

Output Voltage Swing vs. Temperature

200

250

200

250

100

125

150

175

200

150

100

50

100

150

= +5V

= +25

INPUT OFFSET VOLTAGE 

UNITS

BASED ON 600 OP AMPS

OP295 Input Offset (V

) Distribution

UNITS

250

3.2

0.4

150

100

125

175

200

225

2.8

2.4

2.0

1.6

1.2

0.8

BASED ON 600 OP AMPS

= +5V

40

+85

 V

OP295 TCV

Distribution

5.10

4.50

100

4.80

4.60

25

4.70

50

5.00

4.90

TEMPERATURE 

OUTPUT VOLTAGE SWING Volts

= +5V

= 100k

= 2k

= 10k

Output Voltage Swing vs. Temperature

500

300

150

50

100

300

200

250

350

400

450

250

200

150

100

INPUT OFFSET VOLTAGE 

UNITS

= +5V

= +25

BASED ON 1200 OP AMPS

OP495 Input Offset (V

) Distribution

500

3.2

150

0.4

100

300

200

250

350

400

450

2.8

2.4

2.0

1.6

1.2

0.8

 V

UNITS

= +5V

40

+85

BASED ON 1200 OP AMPS

OP495 TCV

Distribution

REV. B

OP295/OP495Typical Characteristics

= +5V

100

25

50

TEMPERATURE 

INPUT BIAS CURRENT nA

Input Bias Current vs. Temperature

= +5V

TEMPERATURE 

100

25

50

15V

SOURCE

SINK

SOURCE

SINK

OUTPUT CURRENT mA

Output Current vs. Temperature

TEMPERATURE 

OPEN-LOOP GAIN V/

= 2k

100

50

100

25

15V

10V

= 10k

= 100k

Open-Loop Gain vs. Temperature

100

25

50

TEMPERATURE 

OPEN-LOOP GAIN V/

= +5V

= +4V

= 100k

= 10k

= 2k

Open-Loop Gain vs. Temperature

100

10mA

1mA

100

100mV

10mV

1mV

LOAD CURRENT

OUTPUT VOLTAGE

TO RAIL

SOURCE

SINK

= +5V

= +25

Output Voltage to Supply Rail vs. Load Current

OP295/OP495

REV. B

APPLICATIONS
Rail-to-Rail Applications Information

The OP295/OP495 has a wide common-mode input range ex-
tending from ground to within about 800 mV of the positive
supply. There is a tendency to use the OP295/OP495 in buffer
applications where the input voltage could exceed the common-
mode input range. This may initially appear to work because of
the high input range and rail-to-rail output range. But above the
common-mode input range the amplifier is, of course, highly
nonlinear. For this reason it is always required that there be
some minimal amount of gain when rail-to-rail output swing is
desired. Based on the input common-mode range this gain
should be at least 1.2.

Low Drop-Out Reference

The OP295/OP495 can be used to gain up a 2.5 V or other low
voltage reference to 4.5 volts for use with high resolution A/D
converters that operate from +5 volt only supplies. The circuit
in Figure 1 will supply up to 10 mA. Its no-load drop-out volt-
age is only 20 mV. This circuit will supply over 3.5 mA with a
+5 volt supply.

16k

OUT

= 4.5V

1 TO 10

0.001

20k

REF43

+5V

1/2

OP295/
OP495

Figure 1. 4.5 Volt, Low Drop-Out Reference

Low Noise, Single Supply Preamplifier

Most single supply op amps are designed to draw low supply
current, at the expense of having higher voltage noise. This
tradeoff may be necessary because the system must be powered
by a battery. However, this condition is worsened because all
circuit resistances tend to be higher; as a result, in addition to
the op amp's voltage noise, Johnson noise (resistor thermal
noise) is also a significant contributor to the total noise of the
system.

The choice of monolithic op amps that combine the characteris-
tics of low noise and single supply operation is rather limited.
Most single supply op amps have noise on the order of 30 nV/

to 60 nV/

and single supply amplifiers with noise below

5 nV/

do not exist.

In order to achieve both low noise and low supply voltage opera-
tion, discrete designs may provide the best solution. The circuit
on Figure 2 uses the OP295/OP495 rail-to-rail amplifier and a
matched PNP transistor pair--the MAT03--to achieve zero-in/
zero-out single supply operation with an input voltage noise of
3.1 nV/

at 100 Hz. R5 and R6 set the gain of 1000, making

this circuit ideal for maximizing dynamic range when amplifying
low level signals in single supply applications. The OP295/OP495
provides rail-to-rail output swings, allowing this circuit to oper-
ate with 0 to 5 volt outputs. Only half of the OP295/OP495 is
used, leaving the other uncommitted op amp for use elsewhere.

MAT- 03

OUT

LED

10k

C2
10

510

1500pF

100

R2
27k

0.1

2N3906

OP295/

OP495

Figure 2. Low Noise Single Supply Preamplifier

The input noise is controlled by the MAT03 transistor pair and
the collector current level. Increasing the collector current re-
duces the voltage noise. This particular circuit was tested with
1.85 mA and 0.5 mA of current. Under these two cases, the in-
put voltage noise was 3.1 nV/

and 10 nV/

, respectively.

The high collector currents do lead to a tradeoff in supply cur-
rent, bias current, and current noise. All of these parameters will
increase with increasing collector current. For example, typically
the MAT03 has an h

= 165. This leads to bias currents of

A and 3

A, respectively. Based on the high bias currents,

this circuit is best suited for applications with low source imped-
ance such as magnetic pickups or low impedance strain gages.
Furthermore, a high source impedance will degrade the noise
performance. For example, a 1 k

resistor generates 4 nV/

of broadband noise, which is already greater than the noise of
the preamp.

The collector current is set by R1 in combination with the LED
and Q2. The LED is a 1.6 V "Zener" that has a temperature co-
efficient close to that of Q2's base-emitter junction, which pro-
vides a constant 1.0 V drop across R1. With R1 equal to 270

the tail current is 3.7 mA and the collector current is half that,
or 1.85 mA. The value of R1 can be altered to adjust the collec-
tor current. Whenever R1 is changed, R3 and R4 should also be
adjusted. To maintain a common-mode input range that in-
cludes ground, the collectors of the Q1 and Q2 should not go
above 0.5 V--otherwise they could saturate. Thus, R3 and R4
have to be small enough to prevent this condition. Their values
and the overall performance for two different values of R1 are
summarized in Table I. Lastly, the potentiometer, R8, is needed
to adjust the offset voltage to null it to zero. Similar perfor-
mance can be obtained using an OP90 as the output amplifier
with a savings of about 185

A of supply current. However, the

output swing will not include the positive rail, and the band-
width will reduce to approximately 250 Hz.

REV. B

OP295/OP495

Table I. Single Supply Low Noise Preamp Performance

= 1.85 mA

= 0.5 mA

270

1.0 k

R3, R4

200

910

@ 100 Hz

3.15 nV/

8.6 nV/

@ 10 Hz

4.2 nV/

10.2 nV/

4.0 mA

1.3 mA

Bandwidth

1 kHz

Closed-Loop Gain

1000

Driving Heavy Loads

The OP295/OP495 is well suited to drive loads by using a
power transistor, Darlington or FET to increase the current to
the load. The ability to swing to either rail can assure that the
device is turned on hard. This results in more power to the load
and an increase in efficiency over using standard op amps with
their limited output swing. Driving power FETs is also possible
with the OP295/OP495 because of its ability to drive capacitive
loads of several hundred picofarads without oscillating.

Without the addition of external transistors the OP295/OP495
can drive loads in excess of

15 mA with

15 or +30 volt

supplies. This drive capability is somewhat decreased at lower
supply voltages. At

5 volt supplies the drive current is

11 mA.

Driving motors or actuators in two directions in a single supply
application is often accomplished using an "H" bridge. The
principle is demonstrated in Figure 3a. From a single +5 volt
supply this driver is capable of driving loads from 0.8 V to 4.2 V
in both directions. Figure 3b shows the voltages at the inverting
and noninverting outputs of the driver. There is a small crossover
glitch that is frequency dependent and would not cause problems

10k

1.67V

10k

2N2222

OUTPUTS

2N2907

+5V

2.5V

Figure 3a. "H" Bridge

100

1ms

Figure 3b. "H" Bridge Outputs

unless this was a low distortion application such as audio. If this
is used to drive inductive loads, be sure to add diode clamps to
protect the bridge from inductive kickback.

Direct Access Arrangement

OP295/OP495 can be used in a single supply Direct Access Ar-
rangement (DAA) as is shown an in Figure 4. This figure shows
a portion of a typical DM capable of operating from a single
+5 volt supply and it may also work on +3 volt supplies with
minor modifications. Amplifiers A2 and A3 are configured so
that the transmit signal TXA is inverted by A2 and is not in-
verted by A3. This arrangement drives the transformer differen-
tially so that the drive to the transformer is effectively doubled
over a single amplifier arrangement. This application takes ad-
vantage of the OP295/OP495's ability to drive capacitive loads,
and to save power in single supply applications.

2.5V REF

20k

750pF

20k

22.1k

0.1

475

3.3k

0.0047

0.1

0.033

20k

37.4k

390pF

RXA

TXA

OP295/
OP495

1:1

Figure 4. Direct Access Arrangement

A Single Supply Instrumentation Amplifier
The OP295/OP495 can be configured as a single supply instru-
mentation amplifier as in Figure 5. For our example, V

REF

is set

equal to

and V

is measured with respect to V

REF

. The in-

put common-mode voltage range includes ground and the out-
put swings to both rails.

V+

REF

100k

20k

100k

+ V

REF

= 5 +

200k

(

)

1/2

OP295/

OP495

1/2

OP295/

OP495

Figure 5. Single Supply Instrumentation Amplifier

Resistor R

sets the gain of the instrumentation amplifier. Mini-

mum gain is 6 (with no R

). All resistors should be matched in

absolute value as well as temperature coefficient to maximize

OP295/OP495

REV. B

common-mode rejection performance and minimize drift. This
instrumentation amplifier can operate from a supply voltage as
low as 3 volts.

A Single Supply RTD Thermometer Amplifier

This RTD amplifier takes advantage of the rail-to-rail swing of
the OP295/OP495 to achieve a high bridge voltage in spite of a
low 5 V supply. The OP295/OP495 amplifier servos a constant
200

A current to the bridge. The return current drops across

the parallel resistors 6.19 k

and the 2.55 M

, developing a

voltage that is servoed to 1.235 V, which is established by the
AD589 bandgap reference. The 3-wire RTD provides an equal
line resistance drop in both 100

legs of the bridge, thus im-

proving the accuracy.

The AMP04 amplifies the differential bridge signal and converts
it to a single-ended output. The gain is set by the series resis-
tance of the 332

resistor plus the 50

potentiometer. The

gain scales the output to produce a 4.5 V full scale. The
0.22

F capacitor to the output provides a 7 Hz low-pass filter

to keep noise at a minimum.

AMP04

0.22

332

26.7k
0.5%

100

0.5%

100

RTD

2.55M

6.19k
1%

AD589

1.235

37.4k

+5V

200

10-TURNS

ZERO ADJ

26.7k

0.5%

4.5V = 450

0V = 0

+5V

1/2

OP295/

OP495

Figure 6. Low Power RTD Amplifier

A Cold Junction Compensated, Battery Powered
Thermocouple Amplifier

The OP295/OP495's 150

A quiescent current per amplifier

consumption makes it useful for battery powered temperature
measuring instruments. The K-type thermocouple terminates
into an isothermal block where the terminated junctions' ambi-
ent temperatures can be continuously monitored and corrected
by summing an equal but opposite thermal EMF to the ampli-
fier, thereby canceling the error introduced by the cold junctions.

0V = 0

5V = 500

4.99k

1.33M

20k

SCALE

ADJUST

24.3k
1%

24.9k

7.15k

1.235V

AD589

24.9k

500

10-TURN

2.1k
1%

475

1.5M

1N914

ISOTHERMAL

BLOCK

COLD
JUNCTIONS

CHROMEL

K-TYPE
THERMOCOUPLE
40.7

ZERO
ADJUST

ALUMEL

OP295/

OP495

Figure 7. Battery Powered, Cold-Junction Compensated
Thermocouple Amplifier

To calibrate, immerse the thermocouple measuring junction in a
0

C ice bath, adjust the 500

Zero Adjust pot to zero volts out.

Then immerse the thermocouple in a 250

C temperature bath

or oven and adjust the Scale Adjust pot for an output voltage of
2.50 V, which is equivalent to 250

C. Within this temperature

range, the K-type thermocouple is quite accurate and produces
a fairly linear transfer characteristic. Accuracy of

C is achiev-

able without linearization.

Even if the battery voltage is allowed to decay to as low as 7 volts,
the rail-to-rail swing allows temperature measurements to
700

C. However, linearization may be necessary for tempera-

tures above 250

C where the thermocouple becomes rather

nonlinear. The circuit draws just under 500

A supply current

from a 9 V battery.

A 5 V Only, 12-Bit DAC That Swings 0 V to 4.095 V

Figure 8 shows a complete voltage output DAC with wide out-
put voltage swing operating off a single +5 V supply. The serial
input 12-bit D/A converter is configured as a voltage output
device with the 1.235 V reference feeding the current output pin
(I

OUT

) of the DAC. The V

REF

which is normally the input now

becomes the output.

The output voltage from the DAC is the binary weighted volt-
age of the reference, which is gained up by the output amplifier
such that the DAC has a 1 mV per bit transfer function.

+5V

R2
41.2k

R3
5k

100k

REF

SRI

CLK

GND

OUT

DAC8043

+5V

DIGITAL

CONTROL

+5V

AD589

R1
17.8k

+1.23V

(4.096V)

4096

TOTAL POWER DISSIPATION = 1.6mW

OP295/

OP495

Figure 8. A 5 Volt 12-Bit DAC with 0 V to +4.095 Output
Swing

420 mA Current Loop Transmitter

Figure 9 shows a self powered 420 mA current loop transmit-
ter. The entire circuit floats up from the single supply (12 V to
36 V) return. The supply current carries the signal within the 4
to 20 mA range. Thus the 4 mA establishes the baseline

220pF

220

REF02

GND

100

2N1711

100k

HP
5082-2800

100

0 + 3V

100k

10-TURN

1.21M

NULL ADJ

SPAN ADJ

182k

10k

10-TURN

+12V

+36V

100

420mA

1/2

OP295/

OP495

Figure 9. 420 mA Current Loop Transmitter

REV. B

OP295/OP495

current budget with which the circuit must operate. This circuit
consumes only 1.4 mA maximum quiescent current, making 2.6
mA of current available to power additional signal conditioning
circuitry or to power a bridge circuit.

A 3 Volt Low-Dropout Linear Voltage Regulator

Figure 10 shows a simple 3 V voltage regulator design. The
regulator can deliver 50 mA load current while allowing a 0.2 V
dropout voltage. The OP295/OP495's rail-to-rail output swing
handily drives the MJE350 pass transistor without requiring spe-
cial drive circuitry. At no load, its output can swing less than the
pass transistor's base-emitter voltage, turning the device nearly
off. At full load, and at low emitter-collector voltages, the tran-
sistor beta tends to decrease. The additional base current is eas-
ily handled by the OP295/OP495 output.

The amplifier servos the output to a constant voltage, which
feeds a portion of the signal to the error amplifier.

Higher output current, to 100 mA, is achievable at a higher
dropout voltage of 3.8 V.

1000pF

43k

44.2k
1%

30.9k
1%

AD589

1.235V

100

< 50mA

MJE 350

5V TO 3.2V

1/2

OP295/

OP495

Figure 10. 3 V Low Dropout Voltage Regulator

Figure 11 shows the regulator's recovery characteristic when its
output underwent a 20 mA to 50 mA step current change.

100

1ms

20mV

50mA

20mA

OUTPUT

STEP

CURRENT
CONTROL

WAVEFORM

Figure 11. Output Step Load Current Recovery

Low-Dropout, 500 mA Voltage Regulator with Fold-Back
Current Limiting

Adding a second amplifier in the regulation loop as shown in
Figure 12 provides an output current monitor as well as fold-
back current limiting protection.

Amplifier A1 provides error amplification for the normal voltage
regulation loop. As long as the output current is less than 1 am-
pere, amplifier A2's output swings to ground, reverse biasing the
diode and effectively taking itself out of the circuit. However, as
the output current exceeds 1 amp, the voltage that develops
across the 0.1

sense resistor forces the amplifier A2's output

to go high, forward-biasing the diode, which in turn closes the

current limit loop. At this point A2's lower output resistance
dominates the drive to the power MOSFET transistor, thereby
effectively removing the A1 voltage regulation loop from the
circuit.

If the output current greater than 1 amp persists, the current
limit loop forces a reduction of current to the load, which causes
a corresponding drop in output voltage. As the output voltage
drops, the current limit threshold also drops fractionally, result-
ing in a decreasing output current as the output voltage de-
creases, to the limit of less than 0.2 A at 1 V output. This
"fold-back" effect reduces the power dissipation considerably
during a short circuit condition, thus making the power supply
far more forgiving in terms of the thermal design requirements.
Small heat sinking on the power MOSFET can be tolerated.

The OP295's rail-to-rail swing exacts higher gate drive to
the power MOSFET, providing a fuller enhancement to the
transistor. The regulator exhibits 0.2 V dropout at 500 mA of
load current. At 1 amp output, the dropout voltage is typically
5.6 volts.

124k

REF43

0.01

100k

210k
1%

205k
1%

45.3k
1%

SENSE

0.1

1/4W

+5V V

(NORM) = 0.5A

(MAX) = 1A

IRF9531

1N4148

2.500V

1/2

OP295/

OP495

1/2

OP295/

OP495

Figure 12. Low Dropout, 500 mA Voltage Regulator with
Fold-Back Current Limiting

Square Wave Oscillator
The circuit in Figure 13 is a square wave oscillator (note the
positive feedback). The rail-to-rail swing of the OP295/OP495
helps maintain a constant oscillation frequency even if the sup-
ply voltage varies considerably. Consider a battery powered sys-
tem where the voltages are not regulated and drop over time.
The rail-to-rail swing ensures that the noninverting input sees
the full V+/2, rather than only a fraction of it.

The constant frequency comes from the fact that the 58.7 k

feedback sets up Schmitt Trigger threshold levels that are di-
rectly proportional to the supply voltage, as are the RC charge
voltage levels. As a result, the RC charge time, and therefore the
frequency, remains constant independent of supply voltage. The
slew rate of the amplifier limits the oscillation frequency to a
maximum of about 800 Hz at a +5 V supply.

Single Supply Differential Speaker Driver

Connected as a differential speaker driver, the OP295/OP495
can deliver a minimum of 10 mA to the load. With a 600

load, the OP295/OP495 can swing close to 5 volts peak-to-peak
across the load.

OP295/OP495

REV. B

58.7k

100k

V+

FREQ OUT

OSC

< 350Hz @ V+ = +5V

100k

1/2

OP295/

OP495

Figure 13. Square Wave Oscillator Has Stable Frequency
Regardless of Supply Changes

10k

V+

10k

100k

90.9k

20k

V+

SPEAKER

2.2

1/4

OP295/

OP495

1/4

OP295/

OP495

1/4

OP295/

OP495

Figure 14. Single Supply Differential Speaker Driver

High Accuracy, Single-Supply, Low Power Comparator

The OP295/OP495 makes an accurate open-loop comparator.
With a single +5 V supply, the offset error is less than 300

V. Fig-

ure 15 shows the OP295/OP495's response time when operating
open-loop with 4 mV overdrive. It exhibits a 4 ms response time at
the rising edge and a 1.5 ms response time at the falling edge.

100

5ms

OUTPUT

(5mV OVERDRIVE

@ OP295 INPUT)

INPUT

Figure 15. Open-Loop Comparator Response Time with
5 mV Overdrive

OP295/OP495 SPICE MODEL Macro-Model

* Node Assignments
*

Noninverting Input

Inverting Input

Positive Supply

Negative Supply

Output

*
*
.SUBCKT OP295

*
* INPUT STAGE
*
I1

2E-6

5E3

CIN

2E-12

IOS

0.5E-9

EOS

POLY (1) (31,39) 30E-6 0.024

25.8E3

*
* GAIN STAGE
*
R7

270E6

POLY (1) (9,8) 4.26712E-9 27.8E-6

EREF 98

(39, 0) 1

100E3

*
* COMMON MODE STAGE
*
ECM

POLY(2) (1,39) (2,39) 0 0.5 0.5

R12

1E6

R13

100

*
* OUTPUT STAGE
*
I2

1.59E-6

DC 2.2763

QNA

1.0

R11

MN L=9E-6 W=102E-6 AD=15E-10 AD=15E-10

MN L=9E-6 W=50E-6 AD=75E-11 AS=75E-11

DC 6

MN L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9

QPA

1.0

QPA

4.0

2.2E6

QPA

1.0

20E-12

MP L=9E-6 W=27E-6 AD=405E-12 AS=405E-12

MP L=9E-6 W=2000E-6 AD=30E-9 AS=30E-9

DC 6

R10

6E3

50E-12

.MODEL QNA NPN (IS=1.19E-16 BF=253 NF=0.99 VAF=193 IKF=2.76E-3
+ ISE=2.57E-13 NE=5 BR=0.4 NR=0.988 VAR=15 IKR=1.465E-4
+ ISC=6.9E-16 NC=0.99 RB=2.0E3 IRB=7.73E-6 RBM=132.8 RE=4
RC=209
+ CJE=2.1E-13 VJE=0.573 MJE=0.364 FC=0.5 CJC=1.64E-13 VJC=0.534
MJC=0.5
+ CJS=1.37E-12 VJS=0.59 MJS=0.5 TF=0.43E-9 PTF=30)
.MODEL QPA PNP (IS=5.21E-17 BF=131 NF=0.99 VAF=62 IKF=8.35E-4
+ ISE=1.09E-14 NE=2.61 BR=0.5 NR=0.984 VAR=15 IKR=3.96E-5
+ ISC=7.58E-16 NC=0.985 RB=1.52E3 IRB=1.67E-5 RBM=368.5 RE=6.31
RC=354.4
+ CJE=1.1E-13 VJE=0.745 MJE=0.33 FC=0.5 CJC=2.37E-13 VJC=0.762
MJC=0.4
+ CJS =7.11E-13 VJS=0.45 MJS=0.412 TF=1.0E-9 PTF=30)
.MODEL MN NMOS (LEVEL=3 VTO=1.3 RS=0.3 RD=0.3
+ TOX=8.5E-8 LD=1.48E-6 NSUB=1.53E16 UO=650 DELTA=10 VMAX=2E5
+ XJ=1.75E-6 KAPPA=0.8 ETA=0.066 THETA=0.01 TPG=1 CJ=2.9E-4
PB=0.837
+ MJ=0.407 CJSW=0.5E-9 MJSW=0.33)
.MODEL MP PMOS (LEVEL=3 VTO=1.1 RS=0.7 RD=0.7
+ TOX=9.5E-8 LD=1.4E-6 NSUB=2.4E15 UO=650 DELTA=5.6 VMAX=1E5
+ XJ=1.75E-6 KAPPA=1.7 ETA=0.71 THETA=5.9E-3 TPG=1 CJ=1.55E-4
PB=0.56
+ MJ=0.442 CJSW=0.4E-9 MJSW=0.33)
.MODEL DX D(IS=1E-15)
.MODEL DZ D (IS=1E-15, BV=7)
.MODEL QP PNP (BF=125)

.ENDS

REV. B

OP295/OP495

C1806a107/95

PRINTED IN U.S.A.

OUTLINE DIMENSIONS

Dimensions shown in inches and (mm)

8 Lead Plastic DIP (P Suffix)

0.160 (4.06)

0.115 (2.93)

0.130

(3.30)

MIN

0.210

(5.33)

MAX

0.015

(0.381) TYP

0.430 (10.92)

0.348 (8.84)

0.280 (7.11)

0.240 (6.10)

0.070 (1.77)

0.045 (1.15)

0.022 (0.558)

0.014 (0.356)

0.325 (8.25)

0.300 (7.62)

- 15

0.100

(2.54)

BSC

0.015 (0.381)

0.008 (0.204)

SEATING

PLANE

0.195 (4.95)

0.115 (2.93)

14-Lead Plastic DIP (P Suffix)

PIN 1

0.280 (7.11)
0.240 (6.10)

0.210

(5.33)

MAX

0.160 (4.06)
0.115 (2.92)

0.795 (20.19)
0.725 (18.42)

0.022 (0.558)

0.014 (0.36)

0.100
(2.54)

BSC

0.070 (1.77)
0.045 (1.15)

SEATING
PLANE

0.130
(3.30)
MIN

0.015
(0.381)
MIN

0.325 (8.25)
0.300 (7.62)

0.015 (0.38)
0.008 (0.20)

8-Lead Narrow-Body SO (S Suffix)

0.0098 (0.25)

0.0075 (0.19)

0.0500 (1.27)

0.0160 (0.41)

0.0196 (0.50)

0.0099 (0.25)

x 45

PIN 1

0.1574 (4.00)

0.1497 (3.80)

0.2440 (6.20)

0.2284 (5.80)

0.0192 (0.49)

0.0138 (0.35)

0.0500

(1.27)

BSC

0.0688 (1.75)

0.0532 (1.35)

0.0098 (0.25)

0.0040 (0.10)

0.1968 (5.00)

0.1890 (4.80)

16-Lead Wide-Body SO (S Suffix)

PIN 1

0.2992 (7.60)

0.2914 (7.40)

0.4193 (10.65)

0.3937 (10.00)

0.0192 (0.49)

0.0138 (0.35)

0.0500 (1.27)

BSC

0.1043 (2.65)

0.0926 (2.35)

0.4133 (10.50)

0.3977 (10.10)

0.0118 (0.30)

0.0040 (0.10)

0.0125 (0.32)

0.0091 (0.23)

0.0500 (1.27)

0.0157 (0.40)

0.0291 (0.74)

0.0098 (0.25)

x 45

Ð­Ð»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: OP295GBC

ÐÐ»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: OP295GBC