FEATURES

GAIN-BANDWIDTH: 1.6GHz

STABLE IN GAINS

LOW DIFFERENTIAL GAIN/PHASE
ERRORS: 0.015%/0.006

HIGH SLEW RATE: 650V/

FAST 12-BIT SETTLING: 18ns (0.01%)

HIGH COMMON-MODE REJECTION: 80dB

LOW HARMONICS: 72dBc at 10MHz

OPA641

Wideband Voltage Feedback

OPERATIONAL AMPLIFIER

APPLICATIONS

COMMUNICATIONS

MEDICAL IMAGING

TEST EQUIPMENT

CCD IMAGING

ADC/DAC GAIN AMPLIFIER

HIGH-RESOLUTION VIDEO

LOW NOISE PREAMPLIFIER

ACTIVE FILTERS

DESCRIPTION

The OPA641 is an extremely wideband operational
amplifier featuring low noise, high slew rate and high
spurious free dynamic range.

The OPA641 is conservatively compensated for sta-
bility in gains of 2 or greater. This amplifier has a fully
symmetrical differential input due to its "classical"

operational amplifier circuit architecture. This allows
the OPA641 to be used in all op amp applications
requiring high speed and precision.

Low noise, wide bandwidth, and high linearity make
this amplifier suitable for a variety of RF, video, and
imaging applications.

Current

Mirror

Output

Stage

Non-Inverting

Input

Inverting

Input

7, 8

4, 5

Output

International Airport Industrial Park · Mailing Address: PO Box 11400 · Tucson, AZ 85734 · Street Address: 6730 S. Tucson Blvd. · Tucson, AZ 85706

Tel: (520) 746-1111 · Twx: 910-952-1111 · Cable: BBRCORP · Telex: 066-6491 · FAX: (520) 889-1510 · Immediate Product Info: (800) 548-6132

1993 Burr-Brown Corporation

PDS-1189B

Printed in U.S.A. July, 1994

OPA641

OFFSET VOLTAGE
Input Offset Voltage

Average Drift

HSQ Grade Over Temperature

Power Supply Rejection (+V

)

4.5 to

5.5V

)

INPUT BIAS CURRENT
Input Bias Current

= 0V

Over Specified Temperature

HSQ Grade Over Temperature

Input Offset Current

= 0V

0.2

1.0

Over Specified Temperature

0.5

2.5

2.0

HSQ Grade Over Temperature

1.2

4.0

NOISE
Input Voltage Noise

Noise Density, f = 100Hz

8.0

nV/

f = 10kHz

2.9

nV/

f = 1MHz

2.8

nV/

f = 1MHz to 500MHz

2.8

nV/

Voltage Noise, BW = 100Hz to 500MHz

Vr ms

Input Bias Current Noise Density

f = 0.1Hz to 20kHz

2.0

pA/

Noise Figure (NF)

= 1k

= 50

INPUT VOLTAGE RANGE
Common-Mode Input Range

2.5

2.85

Over Specified Temperature

2.5

2.75

Common-Mode Rejection

0.5V

INPUT IMPEDANCE
Differential

15 || 1

|| pF

Common-Mode

2 || 1

|| pF

OPEN-LOOP GAIN, DC
Open-Loop Voltage Gain

2V, R

= 100

Over Specified Temperature

2V, R

= 100

FREQUENCY RESPONSE, R

= 402

All Four Power Pins Used

Closed-Loop Bandwidth

Gain = +2V/V

800

MHz

Gain = +5V/V

MHz

Gain = +10V/V

MHz

Slew Rate

(1)

G = +2, 2V Step

650

At Minimum Specified Temperature

G = +2, 2V Step

550

Settling Time: 0.01%

G = +2, 2V Step

0.1%

G = +2, 2V Step

Differential Gain at 3.58MHz, G = +2V/V

= 0V to 1.4V, R

= 150

0.015

Differential Phase at 3.58MHz, G = +2V/V

= 0V to 1.4V, R

= 150

0.006

degrees

Gain Flatness

G = +2

0.1

MHz

Spurious Free Dynamic Range

G = +2, f = 5MHz, V

= 2Vp-p

dBc

G = +2, f = 10MHz, V

= 2Vp-p

dBc

OUTPUT
Voltage Output

No Load

Over Specified Temperature

2.6

3.0

HSQ Grade Over Temperature

2.5

2.8

Voltage Output

= 100

Over Specified Temperature

2.25

2.5

HSQ Grade Over Temperature

2.0

2.3

Current Output

Over Specified Temperature

HSQ Grade Over Temperature

Short Circuit Current

Output Resistance

1MHz, G = +2V/V

0.04

OPA641H, P, U

OPA641HSQ, PB, UB

PARAMETER

CONDITIONS

MIN

TYP

MAX

MIN

TYP

MAX

UNITS

SPECIFICATIONS

ELECTRICAL

= +25

C, V

5V, R

= 100

, C

= 2pF, R

= 402

and all four power supply pins are used unless otherwise noted.

OPA641

POWER SUPPLY
Specified Operating Voltage

MIN

to T

MAX

Operating Voltage Range

MIN

to T

MAX

4.5

5.5

Quiescent Current

Over Specified Temperature

TEMPERATURE RANGE
Specification: H, P, PB, U, UB

Ambient

+85

HSQ

Ambient

+125

Thermal Resistance

, Junction to Ambient

120

C/W

170

C/W

120

C/W

NOTE: (1) Slew rate is rate of change from 10% to 90% of output voltage step.

The information provided herein is believed to be reliable; however, BURR-BROWN assumes no responsibility for inaccuracies or omissions. BURR-BROWN assumes
no responsibility for the use of this information, and all use of such information shall be entirely at the user's own risk. Prices and specifications are subject to change
without notice. No patent rights or licenses to any of the circuits described herein are implied or granted to any third party. BURR-BROWN does not authorize or warrant
any BURR-BROWN product for use in life support devices and/or systems.

SPECIFICATIONS

(CONT)

ELECTRICAL

= +25

C, V

5V, R

= 100

, C

= 2pF, R

= 402

, and all four power supply pins are used unless otherwise noted.

OPA641H, P, U

OPA641HSQ, PB, UB

PARAMETER

CONDITIONS

MIN

TYP

MAX

MIN

TYP

MAX

UNITS

OPA641

Basic Model Number
Package Code

H = 8-pin Sidebraze DIP
P = 8-pin Plastic DIP
U = 8-pin Plastic SOIC

Performance Grade Code

S = 55

C to +125

(1)

or No Letter = 40

C to +85

Reliability Screening

Q = Q-Screened (HSQ Model Only)

ORDERING INFORMATION

(

) (

)

( Q )

ABSOLUTE MAXIMUM RATINGS

Supply ..........................................................................................

5.5VDC

Internal Power Dissipation

(1)

....................... See Applications Information

Differential Input Voltage ............................................................ Total V

Input Voltage Range .................................... See Applications Information
Storage Temperature Range: H, HSQ .......................... 65

C to +150

P, PB, U, UB ................. 40

C to +125

Lead Temperature (soldering, 10s) .............................................. +300

(soldering, SOIC 3s) ....................................... +260

Junction Temperature (T

) ............................................................ +175

NOTE: (1) Packages must be derated based on specified

. Maximum

must be observed.

PACKAGE INFORMATION

PACKAGE DRAWING

MODEL

PACKAGE

NUMBER

(1)

OPA641H, HSQ

8-Pin Cerdip

157

OPA641P, PB

8-Pin DIP

006

OPA641U, UB

8-Pin SOIC

182

NOTE: (1) For detailed drawing and dimension table, please see end of data
sheet, or Appendix D of Burr-Brown IC Data Book.

NOTE: (1) The "B" grade of the SOIC package will be designated with a "B". Refer
to the mechanical section for the location.

PIN CONFIGURATION

Top View

DIP/SOIC

NOTE: (1) Making use of all four power supply pins is highly recommended,
although not required. Using these four pins, instead of just pins 4 and 7, will
lower the effective pin impedance and substantially lower distortion.

(1)

Output

(1)

Inverting Input

Non-Inverting Input

ELECTROSTATIC
DISCHARGE SENSITIVITY

Electrostatic discharge can cause damage ranging from per-
formance degradation to complete device failure. Burr-Brown
Corporation recommends that all integrated circuits be handled
and stored using appropriate ESD protection methods.

ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may
be more susceptible to damage because very small parametric
changes could cause the device not to meet published speci-
fications.

OPA641

APPLICATIONS INFORMATION

DISCUSSION OF PERFORMANCE

The OPA641 provides a level of speed and precision not
previously attainable in monolithic form. Unlike current
feedback amplifiers, the OPA641's design uses a "Classi-
cal" operational amplifier architecture and can therefore
be used in all traditional operational amplifier applica-
tions. While it is true that current feedback amplifiers can
provide wider bandwidth at higher gains, they offer some
disadvantages. The asymmetrical input characteristics of
current feedback amplifiers (i.e., one input is a low imped-
ance) prevents them from being used in a variety of
applications. In addition, unbalanced inputs make input
bias current errors difficult to correct. Bias current cancel-
lation through matching of inverting and non-inverting
input resistors is impossible because the input bias cur-
rents are uncorrelated. Current noise is also asymmetrical
and is usually significantly higher on the inverting input.
Perhaps most important, settling time to 0.01% is often
extremely poor due to internal design tradeoffs. Many
current feedback designs exhibit settling times to 0.01% in
excess of 10 microseconds even though 0.1% settling
times are reasonable. Such amplifiers are completely in-
adequate for fast settling 12-bit applications.

The OPA641's "Classical" operational amplifier architec-
ture employs true differential and fully symmetrical inputs
to eliminate these troublesome problems. All traditional
circuit configurations and op amp theory apply to the
OPA641.

WIRING PRECAUTIONS

Maximizing the OPA641's capability requires some wiring
precautions and high-frequency layout techniques. Oscilla-
tion, ringing, poor bandwidth and settling, gain peaking, and
instability are typical problems plaguing all high-speed
amplifiers when they are improperly used. In general, all
printed circuit board conductors should be wide to provide
low resistance, low impedance signal paths. They should
also be as short as possible. The entire physical circuit
should be as small as practical. Stray capacitances should be
minimized, especially at high impedance nodes, such as the
amplifier's input terminals. Stray signal coupling from the
output or power supplies to the inputs should be minimized.
All circuit element leads should be no longer than 1/4 inch
(6mm) to minimize lead inductance, and low values of
resistance should be used. This will minimize time constants
formed with the circuit capacitances and will eliminate
stray, parasitic circuits.

Grounding is the most important application consideration
for the OPA641, as it is with all high-frequency circuits.
Oscillations at high frequencies can easily occur if good
grounding techniques are not used. A heavy ground plane
(2 oz. copper recommended) should connect all unused
areas on the component side. Good ground planes can
reduce stray signal pickup, provide a low resistance, low
inductance common return path for signal and power, and

can conduct heat from active circuit package pins into
ambient air by convection.

Supply bypassing is extremely critical and must always be
used, especially when driving high current loads. Both
power supply leads should be bypassed to ground as close as
possible to the amplifier pins. Tantalum capacitors (2.2

with very short leads are recommended. A parallel 0.01

ceramic must also be added. Surface mount bypass capaci-
tors will produce excellent results due to their low lead
inductance. Additionally, suppression filters can be used to
isolate noisy supply lines. Properly bypassed and modula-
tion-free power supply lines allow full amplifier output and
optimum settling time performance.

Points to Remember
1) Making use of all four power supply pins will lower the
effective power supply impedance seen by the input and
output stages. This will improve the AC performance in-
cluding lower distortion. The lowest distortion is achieved
when running separated traces to V

and V

. Power supply

bypassing with 0.01

F and 2.2

F surface mount capacitors

on the topside of the PC board is recommended. It is
essential to keep the 0.01

F capacitor very close to the

power supply pins. Refer to the DEM-OPA64x Datasheet
for the recommended layout and component placement.

2) Whenever possible, use surface mount. Don't use point-to-
point wiring as the increase in wiring inductance will be
detrimental to AC performance. However, if it must be used,
very short, direct signal paths are required. The input signal
ground return, the load ground return, and the power supply
common should all be connected to the same physical point to
eliminate ground loops, which can cause unwanted feedback.

3) Surface mount on the PC Board. Good component selec-
tion is essential. Capacitors used in critical locations should
be a low inductance type with a high quality dielectric
material. Likewise, diodes used in critical locations should
be Schottky barrier types, such as HP5082-2835 for fast
recovery and minimum charge storage. Ordinary diodes will
not be suitable in RF circuits.

4) Whenever possible, solder the OPA641 directly into the
PC board without using a socket. Sockets add parasitic
capacitance and inductance, which can seriously degrade
AC performance or produce oscillations.

5) Use a small feedback resistor (usually 25

) in unity-gain

voltage follower applications for the best performance. For
gain configurations, resistors used in feedback networks
should have values of a few hundred ohms for best perfor-
mance. Shunt capacitance problems limit the acceptable
resistance range to about 1k

on the high end and to a value

that is within the amplifier's output drive limits on the low
end. Metal film and carbon resistors will be satisfactory, but
wirewound resistors (even "non-inductive" types) are abso-
lutely unacceptable in high-frequency circuits. Feedback
resistors should be placed directly between the output and
the inverting input on the backside of the PC board. This
placement allows for the shortest feedback path and the
highest bandwidth. See the demonstration board layout at

OPA641

the end of the datasheet. A longer feedback path than this
will decrease the realized bandwidth substantially.

6) Due to the extremely high bandwidth of the OPA641, the
SOIC package is strongly recommended due its low para-
sitic impedance. The parasitic impedance in the PDIP and
CERDIP packages causes the OPA641 to experience about
5dB of gain peaking in unity-gain configurations. This is
compared with virtually no gain peaking in the SOIC pack-
age in unity-gain. The gain peaking in the PDIP and CERDIP
packages is minimized in gains of 4 or greater, however.
Surface mount components (chip resistors, capacitors, etc.)
also have low lead inductance and are therefore strongly
recommended.

7) Avoid overloading the output. Remember that output
current must be provided by the amplifier to drive its own
feedback network as well as to drive its load. Lowest
distortion is achieved with high impedance loads.

8) Don't forget that these amplifiers use

5V supplies.

Although they will operate perfectly well with +5V and
5.2V, use of

15V supplies will destroy the part.

9) Standard commercial test equipment has not been de-
signed to test devices in the OPA641's speed range. Bench-
top op amp testers and ATE systems will require a special
test head to successfully test these amplifiers.

10) Terminate transmission line loads. Unterminated lines,
such as coaxial cable, can appear to the amplifier to be a
capacitive or inductive load. By terminating a transmission
line with its characteristic impedance, the amplifier's load
then appears purely resistive.

11) Plug-in prototype boards and wire-wrap boards will not
be satisfactory. A clean layout using RF techniques is
essential; there are no shortcuts.

OFFSET VOLTAGE ADJUSTMENT

If additional offset adjustment is needed, the circuit in
Figure 1 can be used without degrading offset drift with
temperature. Avoid external adjustment whenever possible

since extraneous noise, such as power supply noise, can be
inadvertently coupled into the amplifier's inverting input
terminal. Remember that additional offset errors can be
created by the amplifier's input bias currents. Whenever
possible, match the impedance seen by both inputs as is
shown with R

This will reduce input bias current errors to

the amplifier's offset current.

INPUT PROTECTION

Static damage has been well recognized for MOSFET de-
vices, but any semiconductor device deserves protection
from this potentially damaging source. The OPA641 incor-
porates on-chip ESD protection diodes as shown in Figure 2.
This eliminates the need for the user to add external protec-
tion diodes, which can add capacitance and degrade AC
performance.

All pins on the OPA641 are internally protected from ESD

NOTE: (1) R

is optional and can be used to cancel offset errors due to input

bias currents.

FIGURE 1. Offset Voltage Trim.

by means of a pair of back-to-back reverse-biased diodes to
either power supply as shown. These diodes will begin to
conduct when the input voltage exceeds either power supply
by about 0.7V. This situation can occur with loss of the
amplifier's power supplies while a signal source is still
present. The diodes can typically withstand a continuous
current of 30mA without destruction. To insure long term
reliability, however, diode current should be externally lim-
ited to 10mA or so whenever possible.

The OPA641 utilizes a fine geometry high speed process
that withstands 500V using Human Body Model and 100V
using the Machine Model. However, static damage can
cause subtle changes in amplifier input characteristics with-
out necessarily destroying the device. In precision opera-
tional amplifiers, this may cause a noticeable degradation of
offset voltage and drift. Therefore, static protection is strongly
recommended when handling the OPA641.

OUTPUT DRIVE CAPABILITY

The OPA641 has been optimized to drive 75

and 100

resistive loads. The device can drive 2Vp-p into a 75

load.

This high-output drive capability makes the OPA641 an
ideal choice for a wide range of RF, IF, and video applica-
tions. In many cases, additional buffer amplifiers are un-
needed.

ESD Protection diodes internally
connected to all pins.

FIGURE 2. Internal ESD Protection.

External

Internal
Circuitry

OPA641

(1)

= R

|| R

Trim

20k

or Ground

Output Trim Range +V

to V

Trim

47k

Trim

10µF

OPA641

Many demanding high-speed applications such as
ADC/DAC buffers require op amps with low wideband
output impedance. For example, low output impedance is
essential when driving the signal-dependent capacitances at
the inputs of flash A/D converters. As shown in Figure 3,
the OPA641 maintains very low closed-loop output imped-
ance over frequency. Closed-loop output impedance in-
creases with frequency since loop gain is decreasing with
frequency.

THERMAL CONSIDERATIONS

The OPA641 does not require a heat sink for operation in
most environments. At extreme temperatures and under full
load conditions a heat sink may be necessary.

The internal power dissipation is given by the equation
P

= P

+ P

, where P

is the quiescent power dissipa-

tion and P

is the power dissipation in the output stage due

to the load. (For

5V, P

= 10V

24mA =

240mW, max). For the case where the amplifier is driving a
grounded load (R

) with a DC voltage (

OUT

) the maxi-

mum value of P

occurs at

OUT

/2, and is equal

to P

, max = (

)

/4R

. Note that it is the voltage across

the output transistor, and not the load, that determines the
power dissipated in the output stage.

The short-circuit condition represents the maximum amount
of internal power dissipation that can be generated. The
variation of output current with temperature is shown in the
Typical Performance Curves.

CAPACITIVE LOADS

The OPA641's output stage has been optimized to drive low
resistive loads. Capacitive loads, however, will decrease the
amplifier's phase margin which may cause high frequency
peaking or oscillations. Capacitive loads greater than 5pF
should be buffered by connecting a small resistance, usually
5

to 25

, in series with the output as shown in Figure 4.

This is particularly important when driving high capacitance
loads such as flash A/D converters.

In general, capacitive loads should be minimized for opti-
mum high frequency performance. Coax lines can be driven
if the cable is properly terminated. The capacitance of coax

cable (29pF/foot for RG-58) will not load the amplifier
when the coaxial cable or transmission line is terminated in
its characteristic impedance.

COMPENSATION

The OPA641 is internally compensated and is stable in unity
gain with a phase margin of approximately 60

. However,

the unity gain buffer is the most demanding circuit configu-
ration for loop stability and oscillations are most likely to
occur in this gain. If possible, use the device in a noise gain
of two or greater to improve phase margin and reduce the
susceptibility to oscillation. (Note that, from a stability
standpoint, an inverting gain of 1V/V is equivalent to a
noise gain of 2.) Gain and phase response for other gains are
shown in the Typical Performance Curves.

The high-frequency response of the OPA641 in a good
layout is very flat with frequency. However, some circuit
configurations such as those where large feedback resis-
tances are used, can produce high-frequency gain peaking.
This peaking can be minimized by connecting a small
capacitor in parallel with the feedback resistor. This capaci-
tor compensates for the closed-loop, high frequency, transfer
function zero that results from the time constant formed by
the input capacitance of the amplifier (typically 2pF after PC
board mounting), and the input and feedback resistors. The
selected compensation capacitor may be a trimmer, a fixed
capacitor, or a planned PC board capacitance. The capaci-
tance value is strongly dependent on circuit layout and
closed-loop gain. Using small resistor values will preserve
the phase margin and avoid peaking by keeping the break
frequency of this zero sufficiently high. When high closed-
loop gains are required, a three-resistor attenuator (tee net-
work) is recommended to avoid using large value resistors
with large time constants.

SETTLING TIME

Settling time is defined as the total time required, from the
input signal step, for the output to settle to within the
specified error band around the final value. This error band is
expressed as a percentage of the value of the output transition,
a 2V step. Thus, settling time to 0.01% requires an error band
of

200

V centered around the final value of 2V.

FIGURE 3. Small-Signal Output Impedance vs Frequency.

FIGURE 4. Driving Capacitive Loads.

OPA641

typically 5

to 25

)

100

10.0

1.0

0.1

0.01

0.001

10k

Frequency (Hz)

Output Impedance (

)

100k

10M

100M

= +2V/V

OPA641

The third-order intercept point is an important parameter for
many RF amplifier applications. Figure 6 shows the
OPA641's single-tone third-order intercept versus frequency.
This curve is particularly useful for determining the magni-
tude of the third harmonic as a function of frequency, load
resistance, and gain. For example, assume that the applica-
tion requires the OPA641 to operate in a gain of +2V/V and
drive 2Vp-p into 100

at a frequency of 5MHz. Referring to

Figure 6 we find that the intercept point is +38dBm. The
magnitude of the third harmonic can now be easily calcu-
lated from the expression:

Third Harmonic (dBc) = 2(OPI

P P

)

where OPI

P = third-order output intercept, dBm

= output level/tone, dBm/tone

For this case OPI

P = 38dBm, P

= 7dBm, and the third

harmonic = 2(38 7) = 62dB below the fundamental tone.
The OPA641's low IMD makes the device an excellent
choice for a variety of RF signal processing applications.
The value for the two-tone third-order intercept is typically
6dB lower than the single-tone value.

Settling time, specified in an inverting gain of one, occurs in
only 18ns to 0.01% for a 2V step, making the OPA641 one
of the fastest settling monolithic amplifiers commercially
available. Settling time increases with closed-loop gain and
output voltage change as described in the Typical Perform-
ance Curves. Preserving settling time requires critical atten-
tion to the details as mentioned under "Wiring Precautions."
The amplifier also recovers quickly from input overloads.
Overload recovery time to linear operation from a 50%
overload is typically only 30ns.

In practice, settling time measurements on the OPA641
prove to be very difficult to perform. Accurate measurement
is next to impossible in all but the very best equipped labs.
Among other things, a fast flat-top generator and high speed
oscilloscope are needed. Unfortunately, fast flat-top genera-
tors, which settle to 0.01% in sufficient time, are scarce and
expensive. Fast oscilloscopes, however, are more commonly
available. For best results, a sampling oscilloscope is recom-
mended. Sampling scopes typically have bandwidths that
are greater than 1GHz and very low capacitance inputs.
They also exhibit faster settling times in response to signals
that would tend to overload a real-time oscilloscope.

Figure 6 shows the test circuit used to measure settling time
for the OPA641. This approach uses a 16-bit sampling
oscilloscope to monitor the input and output pulses. These
waveforms are captured by the sampling scope, averaged,
and then subtracted from each other in software to produce
the error signal. This technique eliminates the need for the
traditional "false-summing junction," which adds extra para-
sitic capacitance. Note that instead of an additional flat-top
generator, this technique uses the scope's built-in calibration
source as the input signal.

DIFFERENTIAL GAIN AND PHASE

Differential Gain (DG) and Differential Phase (DP) are
among the more important specifications for video applica-
tions. DG is defined as the percent change in closed-loop
gain over a specified change in output voltage level. DP is
defined as the change in degrees of the closed-loop phase
over the same output voltage change. Both DG and DP are
specified at the NTSC sub-carrier frequency of 3.58MHz.
DG and DP increase with closed-loop gain and output
voltage transition. All measurements were performed using
a Tektronix model VM700 Video Measurement Set.

DISTORTION AND NOISE

The OPA641's harmonic distortion characteristics vs fre-
quency and power output in the Typical Performance Curves.
Distortion can be further improved by increasing the load
resistance (refer to Figure 5). Remember to include the
contribution of the feedback resistance when calculating the
effective load resistance seen by the amplifier.

Although harmonic distortion may decrease with higher
load resistances (i.e., higher feedback resistors), the effec-
tive output noise will increase due to the higher resistance.
Therefore, noise or harmonic distortion may be optimized
by picking the appropriate feedback resistor.

FIGURE 5. 5MHz Harmonic Distortion vs Load Resistance.

FIGURE 6. Single-Tone Third-Order Intercept Point vs Fre-

quency.

100

Load Resistance (

)

Harmonic Distortion (dBc)

100

10k

G = +2, V

= 2Vp-p, f

= 5MHz

Frequency (Hz)

10M

100M

Third-Order Intercept Point (dBm)

G = +2V/V

OPA641

NOISE FIGURE

The OPA641 voltage and current noise spectral densities are
specified in the Typical Performance Curves. For RF appli-
cations, however, Noise Figure (NF) is often the preferred
noise specification since it allows system noise performance
to be more easily calculated. The OPA641's Noise Figure vs
Source Resistance is shown in Figure 7.

ENVIRONMENTAL (Q) SCREENING

The inherent reliability of a semiconductor device is con-
trolled by the design, materials and fabrication of the device
--it cannot be improved by testing. However, the use of
environmental screening can eliminate the majority of those
units which would fail early in their lifetimes (infant mortal-
ity) through the application of carefully selected accelerated
stress levels. Burr-Brown "Q-Screening" provides environ-
mental screening to our standard industrial products, thus
enhancing reliability. The screening illustrated in the follow-
ing table is performed to selected stress levels similar to
those of MIL-STD-883.

FIGURE 7. Noise Figure vs Source Resistance.

SPICE MODELS

Computer simulation using SPICE is often useful when
analyzing the performance of analog circuits and systems.
This is particularly true for Video and RF amplifier circuits
where parasitic capacitance and inductance can have a major
effect on circuit performance. SPICE models are available
for the OPA641. Contact Burr-Brown Applications Depart-
ment to receive a spice diskette.

SCREEN

METHOD

Internal Visual

Burr-Brown QC4118

Stabilization Bake

Temperature = 150

C, 24 hrs

Temperature Cycling

Temperature = 65

C to 150

C, 10 cycles

Burn-In Test

Temperature = 125

C, 160 hrs minimum

Centrifuge

20,000G

Hermetic Seal

Fine: He leak rate < 5 x 1

atm cc/s, 30PSiG

Gross: per Fluorocarbon bubble test, 60PSiG

Electrical Tests

As described in specifications tables.

External Visual

Burr-Brown QC5150

NOTE: Q-Screening is available on the HSQ package only.

DEMONSTRATION BOARDS

Demonstration boards to speed prototyping are available.
Refer to the DEM-OPA64X Datasheet for details.

Noise Figure (dB)

Source Resistance (

)

100

10k

100k

NF = 10 LOG 1 +

+ (I

)

4KTR

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