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Электронный компонент: FAN5093

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www.fairchildsemi.com
REV. 1.1.0 3/27/03
Pentium is a registered trademark of Intel Corporation. Athlon is a registered trademark of AMD. Programmable Active Droop is a trademark of Fairchild Semiconductor.
Block Diagram
22
VID0
11
10
14
13
17
16
24
-
+
OSC
Digital
Control
Power
Good
5-Bit
DAC
VID1
VID2
VID3
VID4
1 2 3 4 5
+12V
PWRGD
-
+
ENABLE/SS
VO
AGND
7
DROOP/E*
18
6
+12V
15
5V Reg
BYPASS
23
R
T
+12V
+12V
12
8
9
21
+12V
Digital
Control
-
+
-
+
-
+
19
ILIM
20
BOOT A
BOOT B
GNDA
Current
Limit
UVL O
FAN5093
Two Phase Interleaved Synchronous Buck Converter
for VRM 9.x Applications
Features
Programmable output from 1.10V to 1.85V in 25mV steps
using an integrated 5-bit DAC
Two interleaved synchronous phases for maximum
performance
100nsec transient response time
Built-in current sharing between phases
Remote sense
Programmable Active Droop
(Voltage Positioning)
Programmable switching frequency from 100KHz to
1MHz per phase
Adaptive delay gate switching
Integrated high-current gate drivers
Integrated Power Good, OV, UV, Enable/Soft Start
functions
Drives N-channel MOSFETs
Operation optimized for 12V operation
High efficiency mode (E*) at light load
Overcurrent protection using MOSFET sensing
24 pin TSSOP package
Applications
Power supply for Pentium
IV
Power supply for Athlon
VRM for Pentium IV processor
Programmable step-down power supply
Description
The FAN5093 is a synchronous two-phase DC-DC controller
IC which provides a highly accurate, programmable output
voltage for VRM 9.x processors. Two interleaved synchro-
nous buck regulator phases with built-in current sharing
operate 180 out of phase to provide the fast transient
response needed to satisfy high current applications while
minimizing external components.
The FAN5093 features Programmable Active Droop
for
transient response with minimum output capacitance. It has
integrated high-current gate drivers, with adaptive delay gate
switching, eliminating the need for external drive devices.
The FAN5093 uses a 5-bit D/A converter to program the
output voltage from 1.10V to 1.85V in 25mV steps with an
accuracy of 1%. The FAN5093 uses a high level of integra-
tion to deliver load currents in excess of 50A from a 12V
source with minimal external circuitry.
The FAN5093 also offers integrated functions including
Power Good, Output Enable/Soft Start, under-voltage lock-
out, over-voltage protection, and adjustable current limiting
with independent current sense on each phase. It is available
in a 24 pin TSSOP package.
FAN5093
PRODUCT SPECIFICATION
2
REV. 1.1.0 3/27/03
Pin Assignments
Pin Definitions
Pin Number
Pin Name
Pin Function Description
1-5
VID0-4
Voltage Identification Code Inputs.
Open collector/TTL compatible inputs will
program the output voltage over the ranges specified in Table 1. Internally Pulled-
Up.
6
BYPASS
5V Rail.
Bypass this pin with a 0.1
F ceramic capacitor to AGND.
7
AGND
Analog Ground.
Return path for low power analog circuitry. This pin should be
connected to a low impedance system ground plane to minimize ground loops.
8
LDRVB
Low Side FET Driver for B.
Connect this pin to the gate of an N-channel
MOSFET for synchronous operation. The trace from this pin to the MOSFET gate
should optimally be <0.5
"
.
9
PGNDB
Power Ground B.
Return pin for high currents flowing in low-side MOSFET.
Connect directly to low-side MOSFET source.
10
SWB
High side driver source and low side driver drain switching node B.
Gate
drive return for high side MOSFET, and negative input for low-side MOSFET
current sense.
11
HDRVB
High Side FET Driver B.
Connect this pin to the gate of an N-channel MOSFET.
The trace from this pin to the MOSFET gate should optimally be <0.5
"
.
12
BOOTB
Bootstrap B.
Input supply for high-side MOSFET.
13
BOOTA
Bootstrap A.
Input supply for high-side MOSFET.
14
HDRVA
High Side FET Driver A.
Connect this pin to the gate of an N-channel MOSFET.
The trace from this pin to the MOSFET gate should optimally be <0.5
"
.
15
SWA
High side driver source and low side driver drain switching node A.
Gate
drive return for high side MOSFET, and negative input for low-side MOSFET
current sense.
16
PGNDA
Power Ground A.
Return pin for high currents flowing in low-side MOSFET.
Connect directly to low-side MOSFET source.
17
LDRVA
Low Side FET Driver for A.
Connect this pin to the gate of an N-channel
MOSFET for synchronous operation. The trace from this pin to the MOSFET gate
should optimally be <0.5
"
.
18
VCC
VCC.
Internal IC supply. Connect to system 12V supply, and decouple with a 10
resistor and 1
F ceramic capacitor.
19
PWRGD
Power Good Flag.
An open collector output that will be logic LOW if the output
voltage is less than 350mV less than the nominal output voltage setpoint. Power
Good is prevented from going low until the output voltage is out of spec for
500sec.
FAN5093
VID0
VID1
VID2
VID3
VID4
AGND
BYPASS
LDRVB
PGNDB
SWB
HDRVB
BOOTB
VFB
RT
ENABLE/SS
DROOP/E*
ILIM
PWRGD
VCC
LDRVA
PGNDA
SWA
HDRVA
BOOTA
1
2
3
4
5
6
7
8
9
10
11
12
24
23
22
21
20
19
18
17
16
15
14
13
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
3
Absolute Maximum Ratings
(Absolute Maximum Ratings are the values beyond which the device
may be damaged or have it's useful life impaired. Functional operation under these conditions is not implied.)
Thermal Ratings
Recommended Operating Conditions (See Figure 2)
20
ILIM
Current Limit.
A resistor from this pin to ground sets the over current trip level.
21
DROOP/E*
Droop Control/Energy Star Mode Control.
A resistor from this pin to ground
sets the amount of droop by controlling the gain of the current sense amplifier.
When this pin is pulled high to BYPASS, the phase A drivers are turned off for
Energy-star operation.
22
ENABLE/SS
Output Enable/Softstart.
A logic LOW on this pin will disable the output. An
10A internal current source allows for open collector control. This pin also
doubles as soft start.
23
RT
Frequency Set.
A resistor from this pin to ground sets the switching frequency.
24
VFB
Voltage Feedback.
Connect to the desired regulation point at the output of the
converter.
Parameter
Min.
Max.
Unit
Supply Voltage VCC
15
V
Supply Voltages BOOT to PGND
24
V
BOOT to SW
24
V
Voltage Identification Code Inputs, VID0-VID4
6
V
VFB, ENABLE/SS, PWRGD, DROOP/E*
6
V
SWA, SWB to AGND (<1s)
-3
15
V
PGNDA, PGNDB to AGND
-0.5
0.5
V
Gate Drive Current, peak pulse
3
A
Junction Temperature, T
J
-55
150
C
Storage Temperature
-65
150
C
Parameter
Min.
Typ.
Max.
Unit
Lead Soldering Temperature, 10 seconds
300
C
Power Dissipation, P
D
650
mW
Thermal Resistance Junction-to-Case,
JC
16
C/W
Thremal Resistance Junction-to-Ambient,
JA
84
C/W
Parameter
Conditions
Min.
Max.
Units
Output Driver Supply, BOOTA, B
16
22
V
Ambient Operating Temperature
0
70
C
Supply Voltage V
CC
10.8
13.2
V
Pin Number
Pin Name
Pin Function Description
FAN5093
PRODUCT SPECIFICATION
4
REV. 1.1.0 3/27/03
Electrical Specifications
(V
CC
= 12V, VID = [01111] = 1.475V, and T
A
= +25C using circuit in Figure 2, unless otherwise noted.)
The
denotes specifications which apply over the full operating temperature range.
Notes:
1. As measured at the converter's VFB sense point. For motherboard applications, the PCB layout should exhibit no more than
0.5m
trace resistance between the converter's output capacitors and the CPU. Remote sensing should be used for optimal
performance.
2. Using the VFB pin for remote sensing of the converter's output at the load, the converter will be in compliance with VRM 9.x
specification.
Parameter
Conditions
Min.
Typ.
Max.
Units
Input Supply
UVLO Hysteresis
0.5
V
12V UVLO
Rising Edge
8.5
9.5
10.3
V
12V Supply Current
PWM Output Open
20
mA
Internal Voltage Regulator
BYPASS Voltage
4.75
5
5.25
V
BYPASS Capacitor
100
nF
VREF and DAC
Output Voltage
See Table 1
1.100
1.850
V
Initial Voltage Setpoint
1
I
LOAD
= 0A, VID = [01111]
1.460
1.475
1.490
V
Output Temperature Drift
T
A
= 0 to 70C
5
mV
Line Regulation
V
CC
= 11.4V to 12.6V
130
V
Droop
2
I
LOAD
= 69A, R
DROOP
= 13.3k
56
mV
Programmable Droop Range
0
1.25
m
Response Time
V
out
= 10mV
100
nsec
Current Mismatch
R
DS,on
(A) = R
DS,on
(B),
I
LOAD
= 69A, Droop = 1m
5
%
VID Inputs
Input LOW current, VID pins
V
VID
= 0.4V
-60
A
VID V
IH
2.0
V
VID V
IL
0.8
V
Oscillator
Oscillator Frequency
RT = 54.9k
440
500
560
kHz
Oscillator Range
RT = 137.5k
to 13.75 k
200
2000
kHz
Maximum Duty Cycle
RT = 137.5k
90
%
Minimum LDRV on-time
RT = 13.75k
330
nsec
Gate Drive
Gate Drive On-Resistance
Sink & Source
1.0
Output Driver Rise & Fall Time
See Figure 1, C
L
= 3000pF
20
nsec
Enable/Soft Start
Soft Start Current
10
A
Enable Threshold
ON
OFF
1.0
0.4
V
Power Good
PWRGD Threshold
Logic LOW, V
VID
V
PWRGD
85
88
92
%V
OUT
PWRGD Output Voltage
I
sink
= 4mA
0.4
V
PWRGD Delay
High
Low
500
sec
OVP and OTP
Output Overvoltage Detect
2.1
2.2
2.3
V
Over Temperature Shutdown
130
140
150
C
Over Temperature Hysteresis
40
C
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
5
Gate Drive Test Circuit
Figure 1. Output Drive Timing Diagram
Table 1. Output Voltage Programming Codes
Note:
1. 0 = VID pin is tied to GND.
1 = VID pin is pulled up to 5V.
VID4
VID3
VID2
VID1
VID0
V
OUT
to CPU
1
1
1
1
1
OFF
1
1
1
1
0
1.100V
1
1
1
0
1
1.125V
1
1
1
0
0
1.150V
1
1
0
1
1
1.175V
1
1
0
1
0
1.200V
1
1
0
0
1
1.225V
1
1
0
0
0
1.250V
1
0
1
1
1
1.275V
1
0
1
1
0
1.300V
1
0
1
0
1
1.325V
1
0
1
0
0
1.350V
1
0
0
1
1
1.375V
1
0
0
1
0
1.400V
1
0
0
0
1
1.425V
1
0
0
0
0
1.450V
0
1
1
1
1
1.475V
0
1
1
1
0
1.500V
0
1
1
0
1
1.525V
0
1
1
0
0
1.550V
0
1
0
1
1
1.575V
0
1
0
1
0
1.600V
0
1
0
0
1
1.625V
0
1
0
0
0
1.650V
0
0
1
1
1
1.675V
0
0
1
1
0
1.700V
0
0
1
0
1
1.725V
0
0
1
0
0
1.750V
0
0
0
1
1
1.775V
0
0
0
1
0
1.800V
0
0
0
0
1
1.825V
0
0
0
0
0
1.850V
tR
tF
tDT
tDT
HDRV
LDRV
2V
2V
10%
3000pF
V
OUT
2V
90%
90%
2.5V
10%
FAN5093
PRODUCT SPECIFICATION
6
REV. 1.1.0 3/27/03
Typical Operating Characteristics
(V
CC
= 12V, V
OUT
= 1.475V, and T
A
= +25C using circuit in Figure 2, unless otherwise noted.)
EFFICIENCY VS. OUTPUT CURRENT
LOAD CURRENT (A)
EFFICIENCY (%)
HIGH-SIDE GATE DRIVES, RISE / FALL TIME
HIGH-SIDE GATE DRIVES, E*-MODE
ADAPTIVE GATE DELAY
HIGH-SIDE GATE DRIVES, NORMAL OPERATION
LOW-SIDE GATE DRIVES, RISE / FALL TIME
CH1: HDRVB
CH2: HDRVA
40A Load
CH1: HDRVB
40A Load
CH1: HDRVB
CH2: HDRVA
10A Load
CH1: HDRVB
CH2: LDRVB
40A Load
CH1: LDRVB
40A Load
70
75
80
85
90
0
10
20
30
40
50
60
70
80
2 Phase Mode
E* Mode
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
7
Typical Operating Characteristics
(Continued)
DYNAMIC VID CHANGE
CURRENT LIMIT
CURRENT SHARING, 70A LOAD
OUTPUT RIPPLE, 70A LOAD
CURRENT SHARING, 30A LOAD
DROOP VS. R
DROOP
CH1: I
L1
(5A/div)
CH2: I
L2
(5A/div)
CH1: Iin
CH2: Vout
CH1: I
L1
(5A/div)
CH2: I
L2
(5A/div)
CH1: V
OUT
(1.475V-1.575V)
CH2: VID2
40A Load
Rdroop (K
)
Droop (mV/A) (m
)
0.00
0.50
1.00
1.50
2.00
2.50
3.00
0
10
5
15
20
25
30
35
40
R
T
= 49.9K
R
T
= 61.9K
FAN5093
PRODUCT SPECIFICATION
8
REV. 1.1.0 3/27/03
Typical Operating Characteristics
(Continued)
START-UP, 40A LOAD
LOAD TRANSIENT, 12-52A
POWER-DOWN, 40A LOAD
LOAD TRANSIENT, 0-40A
CH1: Iout (20A/div)
CH2: Vout
CH1: Iout (20A/div)
CH2: Vout
CH1: Vout
CH2: Vin
CH1: Vout
CH2: Vin
VOUT TEMPERATURE VARIATION
TEMPERATURE (
C)
1.501
1.500
1.499
1.498
1.497
1.496
1.495
1.494
0
25
70
100
V
OU
T (V)
CLOSED LOOP RESPONSE, 40A LOAD
FREQUENCY (HZ)
GAIN (dB)
PHASE MARGIN (DEG.)
Gain
Phase Margin
-10
0
10
20
30
40
50
100
1000
10000
100000
0
30
60
90
120
150
180
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
9
Application Circuit
Figure 2. Application Circuit for 70A VRM 9.x Desktop Application
Table 2. FAN5093 Application Bill of Materials for Figure 2
Reference
QTY
Description
Manufacturer / Number
U1
1
IC, PWM, FAN5093
Fairchild FAN5093
Q1-8
8
NFET, 30V, 50A, 9m
Fairchild FDD6696
D1, 2, 3
3
DIOS, 40V, 500mA
Fairchild MBR0540
L1, 2
2
IND, 850nH, 30A, 0.9m
Inter-Technical SCTA5022A-R85M
L3
opt
IND, 750nH, 20A, 3.5m
Inter-Technical SC4015-R75M
R1-4, 9
5
4.7
, 5%
R5-8
4
2.2
, 5%
R10
1
10
, 5%
R11
1
10K, 5%
R12
1
75.0K, 1%
R13
1
13.3K, 1%
R14
1
56.2K, 1%
C1-6
6
1.0f, 25V, 10%, X7R
C7-10
4
0.1f, 16V, 10%, X7R
Cin
4
1500f, 16V, 20%,12m
, Aluminum Electrolytic
Rubycon 16MBZ1500M
Cout
8
2200f, 6.3V, 20%, 12m
, Aluminum Electrolytic
Rubycon 6.3MBZ2200M
+12V
C3
R11
R6
+
Cout
C2
Q5
L2
C7
D3
R2
VID3
Q8
R5
VID4
Q2
Q3
Vi n
R1
VID0
R10
C1
VCore
R3
R13
C9
Q6
Q1
C6
R4
R12
R7
Q7
R14
VID1
+
Cin
Vi n
FAN5093
U1
8
11
1
2
3
4
5
24
20
14
22
6
12
9
13
17
7
18
15
10
16
23
21
19
LDRVB
HDRVB
VID0
VID1
VID2
VID3
VID4
VFB
IL IM
HDRVA
ENABLE/SS
BYPASS
BOOTB
PGND B
BOOTA
LDRVA
AGND
VCC
SWA
SWB
PGNDA
RT
DROOP/E*
PWRGD
R9
C8
L1
R8
+5V
L3
C4
Q4
C5
VID2
C10
D2
D1
FAN5093
PRODUCT SPECIFICATION
10
REV. 1.1.0 3/27/03
Application Information
Operation
The FAN5093 Controller
The FAN5093 is a programmable synchronous two-phase
DC-DC controller IC. When designed with the appropriate
external components, the FAN5093 can be configured to
deliver more than 50A of output current, for VRM 9.x
applications. The FAN5093 functions as a fixed frequency
PWM step down regulator, with a high efficiency mode (E*)
at light load.
Main Control Loop
Refer to the FAN5093 Block Diagram on page 1. The
FAN5093 consists of two interleaved synchronous buck con-
verters, implemented with summing-mode control. Each
phase has its own current feedback, and there is a common
voltage feedback.
The two buck converters controlled by the FAN5093 are
interleaved, that is, they run 180
out of phase. This mini-
mizes the RMS input ripple current, minimizing the number
of input capacitors required. It also doubles the effective
switching frequency, improving transient response.
The FAN5093 implements "summing mode control", which
is different from both classical voltage-mode and current-
mode control. It provides superior performance to either by
allowing a large converter bandwidth over a wide range of
output loads and external components. No external compen-
sation is required.
The control loop of the regulator contains two main sections:
the analog control block and the digital control block. The
analog section consists of signal conditioning amplifiers
feeding into a comparator which provides the input to the
digital control block. The signal conditioning section accepts
inputs from a current sensor and a voltage sensor, with the
voltage sensor being common to both phases, and the current
sensor separate for each. The voltage sensor amplifies the
difference between the VFB signal and the reference voltage
from the DAC and presents the output to each of the two
comparators. The current control path for each phase takes
the difference between its PGND and SW pins when the low-
side MOSFET is on, reproducing the voltage across the
MOSFET and thus the input current; it presents the resulting
signal to the same input of its summing amplifier, adding its
signal to the voltage amplifier's with a certain gain. These
two signals are thus summed together. This sum is then pre-
sented to a comparator looking at the oscillator ramp, which
provides the main PWM control signal to the digital control
block. The oscillator ramps are 180
out of phase with each
other, so that the two phases are on alternately.
The digital control block takes the analog comparator input
to provide the appropriate pulses to the HDRV and LDRV
output pins for each phase. These outputs control the external
power MOSFETs.
Response Time
The FAN5093 utilizes leading-edge, not trailing-edge
control. Conventional trailing-edge control turns on the
high-side MOSFET at a clock signal, and then turns it off
when the error amplifier output voltage is equal to the ramp
voltage. As a result, the response time of a trailing-edge
converter can be as long as the off-time of the high-side
driver, nearly an entire switching period. The FAN5093's
leading-edge control turns the high-side MOSFET on when
the error amplifier output voltage is equal to the ramp volt-
age, and turns it off at the clock signal. As a result, when a
transient occurs, the FAN5093 responds immediately by
turning on the high-side MOSFET. Response time is set by
the internal propagation delays, typically 100nsec. In worst
case, the response time is set by the minimum on-time of the
low-side MOSFET, 330nsec.
Oscillator
The FAN5093 oscillator section runs at a frequency deter-
mined by a resistor from the RT pin to ground according to
the formula
The oscillator generates two internal sawtooth ramps, each at
one-half the oscillator frequency, and running 180
out of
phase with each other. These ramps cause the turn-on time of
the two phases to be phased apart. The oscillator frequency
of the FAN5093 can be programmed from 200KHz to 2MHz
with each phase running at 100KHz to 1MHz, respectively.
Selection of a frequency will depend on various system
performance criteria, with higher frequency resulting in
smaller components but typically lower efficiency.
Remote Voltage Sense
The FAN5093 has true remote voltage sense capability, elim-
inating errors due to trace resistance. To utilize remote sense,
the VFB and AGND pins should be connected as a Kelvin
trace pair to the point of regulation, such as the processor
pins. The converter will maintain the voltage in regulation at
that point. Care is required in layout of these grounds; see the
layout guidelines in this datasheet.
High Current Output Drivers
The FAN5093 contains four high current output drivers that
utilize MOSFETs in a push-pull configuration. The drivers
for the high-side MOSFETs use the BOOT pin for input
power and the SW pin for return. The drivers for the low-side
MOSFETs use the VCC pin for input power and the PGND
pin for return. Typically, the BOOT pin will use a charge
pump as shown in Figure 2. Note that the BOOT and VCC
pins are separated from the chip's internal power and ground,
BYPASS and AGND, for switching noise immunity.
R
T
( )
27.5E9
f Hz
(
)
-------------------
=
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
11
Adaptive Delay Gate Drive
The FAN5093 embodies an advanced design that ensures
minimum MOSFET transition times while eliminating
shoot-through current. It senses the state of the MOSFETs
and adjusts the gate drive adaptively to ensure that they are
never on simultaneously. When the high-side MOSFET turns
off, the voltage on its source begins to fall. When the voltage
there reaches approximately 2.5V, the low-side MOSFETs
gate drive is applied. When the low-side MOSFET turns off,
the voltage at the LDRV pin is sensed. When it drops below
approximately 2V, the high-side MOSFET's gate drive is
applied.
Maximum Duty Cycle
In order to ensure that the current-sensing and charge-
pumping work, the FAN5093 guarantees that the low-side
MOSFET will be on a certain portion of each period. For low
frequencies, this occurs as a maximum duty cycle of approxi-
mately 90%. Thus at 250KHz, with a period of 4sec, the
low-side will be on at least 4sec 10% = 400nsec. At higher
frequencies, this time might fall so low as to be ineffective.
The FAN5093 guarantees a minimum low-side on-time of
approximately 330nsec, regardless of duty cycle.
Current Sensing
The FAN5093 has two independent current sensors, one for
each phase. Current sensing is accomplished by measuring
the source-to-drain voltage of the low-side MOSFET during
its on-time. Each phase has its own power ground pin, to per-
mit the phases to be placed in different locations without
affecting measurement accuracy. For best results, it is impor-
tant to connect the PGND and SW pins for each phase as a
Kelvin trace pair directly to the source and drain, respec-
tively, of the appropriate low-side MOSFET. Care is required
in the layout of these grounds; see the layout guidelines in
this datasheet.
Current Sharing
The two independent current sensors of the FAN5093 operate
with their independent current control loops to guarantee that
the two phases each deliver half of the total output current.
The only mismatch between the two phases occurs if there is
a mismatch between the R
DS,on
of the low-side MOSFETs.
Light Load Efficiency
At light load, the FAN5093 uses a number of techniques to
improve efficiency. Because a synchronous buck converter is
two quadrant, able to both source and sink current, during
light load the inductor current will flow away from the out-
put and towards the input during a portion of the switching
cycle. This reverse current flow is detected by the FAN5093
as a positive voltage appearing on the low-side MOSFET
during its on-time. When reverse current flow is detected,
the low-side MOSFET is turned off for the rest of the cycle,
and the current instead flows through the body diode of the
high-side MOSFET, returning the power to the source. This
technique substantially enhances light load efficiency.
Short Circuit Current Characteristics
(ILIM Pin)
The FAN5093 short circuit current characteristic includes a
function that protects the DC-DC converter from damage in
the event of a short circuit. The short circuit limit is set with
the R
S
resistor, as given by the formula
with I
SC
the desired output current limit, RT the oscillator
resistor and R
DS,on
one phase's low-side MOSFET's on
resistance. Remember to make the R
S
large enough to
include the effects of initial tolerance and temperature varia-
tion on the MOSFETs' R
DS,on
.
Important Note! The oscillator frequency must be selected
before selecting the current limit resistor, because the value
of RT is used in the calculation of R
S
.
When an overcurrent is detected, the high-side MOSFETs
are turned off, and the low-side MOSFETs are turned on,
and they remain in this state until the measured current
through the low-side MOSFET has returned to zero amps.
After reaching zero, the FAN5093 re-soft-starts, ensuring
that it can also safely turn on into a short.
A limitation on the current sense circuit is that I
SC
R
DS,on
must be less that 375mV. To ensure correct operation, use
I
SC
R
DS,on
300mV; between 300mV and 375mV, there
will be some non-linearity in the short-circuit current not
accounted for in the equation.
As an example, consider the typical characteristic of the
DC-DC converter circuit with two FDP6670AL low-side
MOSFETs (R
DS
= 6.5m
maximum at 25C 1.2 at 75C
= 7.8m
each, or 3.9m total) in each phase, RT = 42.1K
(600KHz oscillator) and a 50K
R
S
.
The converter exhibits a normal load regulation characteris-
tic until the voltage across the MOSFETs exceeds the inter-
nal short circuit threshold of 50K
/(3.9m 41.2K 6.66)
= 47A. [Note that this current limit level can be as high as
50K
/(3.5m 41.2K 6.66) = 52A, if the MOSFETs
have typical R
DS,on
rather than maximum, and are at 25
C.]
At this point, the internal comparator trips and signals the
controller to leave on the low-side MOSFETs and keep off
the high-side MOSFETs. The inductor current decreases,
and power is not applied again until the inductor current
reaches 0A and the converter attempts to re-softstart.
E*-mode
In addition, further enhancement in efficiency can be
obtained by putting the FAN5093 into E*-mode. When the
Droop pin is pulled to the 5V BYPASS voltage, the "A"
phase of the FAN5093 is completly turned off, reducing in
half the amount of gate charge power being consumed.
E*-mode can be implemented with the circuit shown in
Figure 3.
R
S
( )
I
SC
R
DS on
,
R
T
3.33
=
FAN5093
PRODUCT SPECIFICATION
12
REV. 1.1.0 3/27/03
Figure 3. Implementing E*-mode Control
Note: The charge pump for the HIDRVs should be based on
the "B" phase of the FAN5093, since the "A" phase is off in
E*-mode.
Internal Voltage Reference
The reference included in the FAN5093 is a precision band-
gap voltage reference. Its internal resistors are precisely
trimmed to provide a near zero temperature coefficient (TC).
Based on the reference is the output from an integrated 5-bit
DAC. The DAC monitors the 5 voltage identification pins,
VID0-4, and scales the reference voltage from 1.100V to
1.850V in 25mV steps.
BYPASS Reference
The internal logic of the FAN5093 runs on 5V. To permit the
IC to run with 12V only, it produces 5V internally with a
linear regulator, whose output is present on the BYPASS pin.
This pin should be bypassed with a 100nF capacitor for noise
suppression. The BYPASS pin should not have any external
load attached to it.
Dynamic Voltage Adjustment
The FAN5093 can have its output voltage dynamically
adjusted to accommodate low power modes. The designer
must ensure that the transitions on the VID lines all occur
simultaneously (within less than
500nsec) to avoid false codes
generating undesired output voltages. The Power Good flag
tracks the VID codes, but has a 500sec delay transitioning
from high to low; this is long enough to ensure that there will
not be any glitches during dynamic voltage adjustment.
Power Good (PWRGD)
The FAN5093 Power Good function is designed in accor-
dance with the Pentium IV DC-DC converter specifications
and provides a continuous voltage monitor on the VFB pin.
The circuit compares the VFB signal to the VREF voltage
and outputs an active-low interrupt signal to the CPU should
the power supply voltage deviate more than -12% of its nom-
inal setpoint. The Power Good flag provides no control func-
tions to the FAN5093.
Output Enable/Soft Start (ENABLE/SS)
The FAN5093 will accept an open collector/TTL signal for
controlling the output voltage. The low state disables the
output voltage. When disabled, the PWRGD output is in the
low state.
Even if an enable is not required in the circuit, this pin
should have attached a capacitor (typically 100nF) to soft-
start the switching. A softstart capacitor may be approxi-
mately chosen by the formula:
where: t
D
is the delay time before the output starts to ramp
t
R
is the ramp time of the output
C
SS
= softstart cap
V
OUT
= nominal output voltage
However, C must be
100nF.
Programmable Active DroopTM
The FAN5093 features Programmable Active DroopTM: as
the output current increases, the output voltage drops propor-
tionately an amount that can be programmed with an exter-
nal resistor. This feature is offered in order to allow
maximum headroom for transient response of the converter.
The current is sensed losslessly by measuring the voltage
across the low-side MOSFET during its on time. Consult the
section on current sensing for details. The droop is adjusted
by the droop resistor changing the gain of the current loop.
Note that this method makes the droop dependent on the
temperature and initial tolerance of the MOSFET, and the
droop must be calculated taking account of these tolerances.
Given a maximum output current, the amount of droop can
be programmed with a resistor to ground on the droop pin,
according to the formula
with V
Droop
the desired droop voltage, RT the oscillator
resistor, I
max
the output current at which the droop is desired,
and R
DS, on
the on-state resistance of one phase's low-side
MOSFET.
Important Note! The oscillator frequency must be selected
before selecting the droop resistor, because the value of R
T
is
used in the calculation of R
Droop
.
Over-Voltage Protection
The FAN5093 constantly monitors the output voltage for
protection against over-voltage conditions. If the voltage at
FAN5098, Pin 6
(Bypass)
FAN5098, Pin 21
(Droop, E*)
2N3906
HI=E*MODE
2N3904
1K
10K
10K
10K
R
DROOP
t
D
C
SS
10
A
--------------
1.7
0.9074
V
OUT
+
(
)
2.5
----------------------------------------------------------
=
t
R
C
SS
10
A
--------------
V
OUT
0.9
V
IN
----------------------------
=
R
Droop
( )
V
Droop
R
T
I
max
R
DS on
,
-------------------------------------
=
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
13
the VFB pin exceeds 2.2V, an over-voltage condition is
assumed and the FAN5093 latches on the external low-side
MOSFET and latches off the high-side MOSFET. The
DC-DC converter returns to normal operation only after V
CC
has been recycled.
Over Temperature Protection
If the FAN5093 die temperature exceeds approximately
150
C, the IC shuts itself off. It remains off until the temper-
ature has dropped approximately 25
C, at which time it
resumes normal operation.
Component Selection
MOSFET Selection
This application requires N-channel Enhancement Mode Field
Effect Transistors. Desired characteristics are as follows:
Low Drain-Source On-Resistance,
R
DS,ON
< 10m
(lower is better);
Power package with low Thermal Resistance;
Drain-Source voltage rating > 15V;
Low gate charge, especially for higher frequency
operation.
For the low-side MOSFET, the on-resistance (R
DS,ON
) is the
primary parameter for selection. Because of the small duty
cycle of the high-side, the on-resistance determines the
power dissipation in the low-side MOSFET and therefore
significantly affects the efficiency of the DC-DC converter.
For high current applications, it may be necessary to use two
MOSFETs in parallel for the low-side for each phase.
For the high-side MOSFET, the gate charge is as important
as the on-resistance, especially with a 12V input and with
higher switching frequencies. This is because the speed of
the transition greatly affects the power dissipation. It may be
a good trade-off to select a MOSFET with a somewhat
higher R
DS,on
, if by so doing a much smaller gate charge is
available. For high current applications, it may be necessary
to use two MOSFETs in parallel for the high-side for each
phase.
At the FAN5093's highest operating frequencies, it may be
necessary to limit the total gate charge of both the high-side
and low-side MOSFETs together, to avert excess power dis-
sipation in the IC.
For details and a spreadsheet on MOSFET selection, refer to
Applications Bulletin AB-8.
Gate Resistors
Use of a gate resistor on every MOSFET is mandatory. The
gate resistor prevents high-frequency oscillations caused by
the trace inductance ringing with the MOSFET gate
capacitance. The gate resistors should be located physically
as close to the MOSFET gate as possible.
The gate resistor also limits the power dissipation inside the
IC, which could otherwise be a limiting factor on the switch-
ing frequency. It may thus carry significant power, especially
at higher frequencies. As an example: The FDB7045L has a
maximum gate charge of 70nC at 5V, and an input capaci-
tance of 5.4nF. The total energy used in powering the gate
during one cycle is the energy needed to get it up to 5V, plus
the energy to get it up to 12V:
This power is dissipated every cycle, and is divided between
the internal resistance of the FAN5093 gate driver and the
gate resistor. Thus,
and each gate resistor thus requires a 1/4W resistor to ensure
worst case power dissipation.
Inductor Selection
Choosing the value of the inductor is a tradeoff between
allowable ripple voltage and required transient response.
A smaller inductor produces greater ripple while producing
better transient response. In any case, the minimum induc-
tance is determined by the allowable ripple. The first order
equation (close approximation) for minimum inductance for
a two-phase converter is:
where:
Vin = Input Power Supply
Vout = Output Voltage
f = DC/DC converter switching frequency
ESR = Equivalent series resistance of all output capacitors in
parallel
Vripple = Maximum peak to peak output ripple voltage
budget.
Schottky Diode Selection
The application circuit of Figure 2 shows a Schottky diode,
D1 (D2 respectively), one in each phase. They are used as
free-wheeling diodes to ensure that the body-diodes in the
low-side MOSFETs do not conduct when the upper
MOSFET is turning off and the lower MOSFETs are turning
on. It is undesirable for this diode to conduct because its high
forward voltage drop and long reverse recovery time
degrades efficiency, and so the Schottky provides a shunt
path for the current. Since this time duration is extremely
short, being minimized by the adaptive gate delay, the
selection criterion for the diode is that the forward voltage of
E
QV
1
2
---C
+
V2
70nC
5V
1
2
---
+
5.4nF
12V
5V
(
)
2
=
=
482nJ
=
P
Rgate
E f
R
gate
R
gate
R
internal
+
(
)
-------------------------------------------------
482nJ
300KHz
=
=
4.7
4.7
0.5
+
---------------------------------
131mW
=
L
min
V
in
2
V
out
f
-----------------------------------
V
out
V
in
-----------
ESR
V
ripple
-----------------
=
FAN5093
PRODUCT SPECIFICATION
14
REV. 1.1.0 3/27/03
the Schottky at the output current should be less than the for-
ward voltage of the MOSFET's body diode. Power capability
is not a criterion for this device, as its dissipation is very
small.
Output Filter Capacitors
The output bulk capacitors of a converter help determine its
output ripple voltage and its transient response. It has
already been seen in the section on selecting an inductor that
the ESR helps set the minimum inductance. For most con-
verters, the number of capacitors required is determined by
the transient response and the output ripple voltage, and
these are determined by the ESR and not the capacitance
value. That is, in order to achieve the necessary ESR to meet
the transient and ripple requirements, the capacitance value
required is already very large.
The most commonly used choice for output bulk capacitors
is aluminum electrolytics, because of their low cost and low
ESR. The only type of aluminum capacitor used should be
those that have an ESR rated at 100kHz. Consult Application
Bulletin AB-14 for detailed information on output capacitor
selection.
For higher frequency applications, particularly those running
the FAN5093 oscillator at >1MHz, Oscon or ceramic capaci-
tors may be considered. They have much smaller ESR than
comparable electrolytics, but also much smaller capacitance.
The output capacitance should also include a number of
small value ceramic capacitors placed as close as possible to
the processor; 0.1F and 0.01F are recommended values.
Input Filter
The DC-DC converter design may include an input inductor
between the system main supply and the converter input as
shown in Figure 2. This inductor serves to isolate the main
supply from the noise in the switching portion of the DC-DC
converter, and to limit the inrush current into the input capac-
itors during power up. A value of 1.3H is recommended.
It is necessary to have some low ESR capacitors at the input
to the converter. These capacitors deliver current when the
high side MOSFET switches on. Because of the interleaving,
the number of such capacitors required is greatly reduced
from that required for a single-phase buck converter. Figure
2 shows 3 x 1500
F, but the exact number required will vary
with the output voltage and current, according to the formula
for the two phase FAN5093, where DC is the duty cycle,
DC = Vout / Vin. Capacitor ripple current rating is a function
of temperature, and so the manufacturer should be contacted
to find out the ripple current rating at the expected opera-
tional temperature. For details on the design of an input filter,
refer to Applications Bulletin AB-16.
Figure 4. Input Filter
Design Considerations and Component
Selection
Additional information on design and component selection
may be found in Fairchild's Application Note 59.
PCB Layout Guidelines
Placement of the MOSFETs relative to the FAN5093 is
critical. Place the MOSFETs such that the trace length of
the HIDRV and LODRV pins of the FAN5093 to the FET
gates is minimized. A long lead length on these pins will
cause high amounts of ringing due to the inductance of the
trace and the gate capacitance of the FET. This noise
radiates throughout the board, and, because it is switching
at such a high voltage and frequency, it is very difficult to
suppress.
In general, all of the noisy switching lines should be kept
away from the quiet analog section of the FAN5093. That
is, traces that connect to pins 8-17 (LODRV, HIDRV,
PGND and BOOT) should be kept far away from the
traces that connect to pins 1 through 7, and pins 18-24.
Place the 0.1F decoupling capacitors as close to the
FAN5093 pins as possible. Extra lead length on these
reduces their ability to suppress noise.
Each power and ground pin should have its own via to the
appropriate plane. This helps provide isolation between
pins.
Place the MOSFETs, inductor, and Schottky of a given
phase as close together as possible for the same reasons as
in the first bullet above. Place the input bulk capacitors as
close to the drains of the high side MOSFETs as possible.
In addition, placement of a 0.1
F decoupling cap right on
the drain of each high side MOSFET helps to suppress
some of the high frequency switching noise on the input
of the DC-DC converter.
Place the output bulk capacitors as close to the CPU as
possible to optimize their ability to supply instantaneous
current to the load in the event of a current transient.
Additional space between the output capacitors and the
CPU will allow the parasitic resistance of the board traces
to degrade the DC-DC converter's performance under
severe load transient conditions, causing higher voltage
deviation. For more detailed information regarding
capacitor placement, refer to Application Bulletin AB-5.
A PC Board Layout Checklist is available from Fairchild
Applications. Ask for Application Bulletin AB-11.
I
rms
I
out
2
--------- 2DC
4DC
2
=
L3
+12V
1000
F, 16V
Electrolytic
Vin
PRODUCT SPECIFICATION
FAN5093
REV. 1.1.0 3/27/03
15
PC Motherboard Sample Layout and Gerber File
A reference design for motherboard implementation of the
FAN5093 along with the PCAD layout Gerber file and silk
screen can be obtained through your local Fairchild repre-
sentative.
FAN5093 Evaluation Board
Fairchild provides an evaluation board to verify the system
level performance of the FAN5093. It serves as a guide to
performance expectations when using the supplied external
components and PCB layout. Please contact your local
Fairchild representative for an evaluation board.
Additional Information
For additional information contact your local Fairchild
representative.
FAN5093
PRODUCT SPECIFICATION
16
REV. 1.1.0 3/27/03
Mechanical Dimensions 24 Lead TSSOP
A
--
.047
--
1.20
Symbol
Inches
Min.
Max.
Min.
Max.
Millimeters
Notes
A1
.002
.006
0.05
0.15
.012
0.30
B
.007
0.19
C
.004
.008
0.09
0.20
E
.169
.177
4.30
4.50
.018
.030
0.45
0.75
.026 BSC
0.65 BSC
e
.252 BSC
6.40 BSC
H
L
0
8
0
8
3
5
2
2
N
24
24

ccc
.004
0.10
--
--
D
.303
.316
7.70
7.90
Notes:
1.
2.
3.
4.
5.
Dimensioning and tolerancing per ANSI Y14.5M-1982.
"D" and "E" do not include mold flash. Mold flash or
protrusions shall not exceed .006 inch (0.15mm).
"L" is the length of terminal for soldering to a substrate.
Terminal numbers are shown for reference only.
Symbol "N" is the maximum number of terminals.
H
E
A
D
e
B
A1
C
ccc C
LEAD COPLANARITY
SEATING
PLANE
L
C
FAN5093
PRODUCT SPECIFICATION
3/27/03 0.0m 005
Stock#DS30005093
2003 Fairchild Semiconductor Corporation
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OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
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Ordering Information
Product Number
Description
Package
FAN5093MTC
VRM 9.x DC-DC Controller
24 pin TSSOP
FAN5093MTCX
VRM 9.x DC-DC Controller
24 pin TSSOP in Tape and Reel