1
LTC1562
Very Low Noise, Low Distortion
Active RC Quad Universal Filter
FEATURES
DESCRIPTIO
N
U
s
Continuous Time--No Clock
s
Four 2nd Order Filter Sections, 10kHz to 150kHz
Center Frequency
s
0.5% Typical Center Frequency Accuracy
s
0.3% Typical Center Frequency Accuracy (A Grade)
s
Wide Variety of Response Shapes
s
Lowpass, Bandpass and Highpass Responses
s
103dB Typical S/N,
5V Supply (Q = 1)
s
97dB Typical S/N, Single 5V Supply (Q = 1)
s
96dB Typical S/(N + THD) at
5V Supply, 20kHz Input
s
Rail-to-Rail Input and Output Voltages
s
DC Accurate to 3mV (Typ)
s
"Zero-Power" Shutdown Mode
s
Single or Dual Supply, 5V to 10V Total
s
Resistor-Programmable f
O
, Q, Gain
The LTC
1562 is a low noise, low distortion continuous-time
filter with rail-to-rail inputs and outputs, optimized for a
center frequency (f
O
) of 10kHz to 150kHz. Unlike most
monolithic filters, no clock is needed. Four independent 2nd
order filter blocks can be cascaded in any combination, such
as one 8th order or two 4th order filters. Each block's
response is programmed with three external resistors for
center frequency, Q and gain, using simple design formulas.
Each 2nd order block provides lowpass and bandpass out-
puts. Highpass response is available if an external capacitor
replaces one of the resistors. Allpass, notch and elliptic
responses can also be realized.
The LTC1562 is designed for applications where dynamic
range is important. For example, by cascading 2nd order
sections in pairs, the user can configure the IC as a dual 4th
order Butterworth lowpass filter with 94dB signal-to-noise
ratio from a single 5V power supply. Low level signals can
exploit the built-in gain capability of the LTC1562. Varying the
gain of a section can achieve a dynamic range as high as
118dB with a
5V supply.
Other cutoff frequency ranges can be provided upon request.
Please contact LTC Marketing.
FREQUENCY (Hz)
10k
GAIN (dB)
10
0
10
20
30
40
50
60
70
80
100k
1M
1562 TA03b
Amplitude Response
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
,
10k
R
IN4
,
10k
R
IN1
10k
V
IN2
V
IN1
SCHEMATIC INCLUDES PIN
NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN)
ALSO CONNECT TO V
SEE TYPICAL APPLICATIONS
FOR OTHER CUTOFF FREQUENCIES
DC ACCURATE, NONINVERTING,
UNITY-GAIN, RAIL-TO-RAIL
INPUT AND OUTPUTS. PEAK
SNR
100dB WITH
5V SUPPLIES
V
OUT1
1562 TA01
V
OUT2
R
IN3
10k
5V
5V
R
Q1
, 5.62k
R21, 10k
R23, 10k
0.1
F
0.1
F
R
Q3
, 5.62k
R24, 10k
R
Q4
, 13k
R
Q2
, 13k
R22, 10k
TYPICAL APPLICATIO
N
U
Dual 4th Order 100kHz Butterworth Lowpass Filter
APPLICATIO
N
S
U
s
High Resolution Systems (14 Bits to 18 Bits)
s
Antialiasing/Reconstruction Filters
s
Data Communications, Equalizers
s
Dual or I-and-Q Channels (Two Matched 4th Order
Filters in One Package)
s
Linear Phase Filtering
s
Replacing LC Filter Modules
, LTC and LT are registered trademarks of Linear Technology Corporation.
2
LTC1562
ABSOLUTE
M
AXI
M
U
M
RATINGS
W
W
W
U
PACKAGE/ORDER I
N
FOR
M
ATIO
N
W
U
U
ORDER PART
NUMBER
LTC1562CG
LTC1562ACG
LTC1562IG
LTC1562AIG
TOP VIEW
G PACKAGE
20-LEAD PLASTIC SSOP
*G PACKAGE PINS 4, 7, 14, 17 ARE
SUBSTRATE/SHIELD CONNECTIONS
AND MUST BE TIED TO V
1
2
3
4
5
6
7
8
9
10
20
19
18
17
16
15
14
13
12
11
INV B
V1 B
V2 B
V
*
V
+
SHDN
V
*
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
*
V
AGND
V
*
V2 D
V1 D
INV D
T
JMAX
= 150
C,
JA
= 136
C/W
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
V
S
=
5V, outputs unloaded, T
A
= 25
C, SHDN pin to logic "low",
unless otherwise noted. AC specs are for a single 2nd order section, R
IN
= R2 = R
Q
=10k
0.1%, f
O
= 100kHz, unless noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
V
S
Total Supply Voltage
4.75
10.5
V
I
S
Supply Current
V
S
=
2.375V, R
L
= 5k, C
L
= 30pF, Outputs at 0V
17.3
19.5
mA
V
S
=
5V, R
L
= 5k, C
L
= 30pF, Outputs at 0V
19
21.5
mA
V
S
=
2.375V, R
L
= 5k, C
L
= 30pF, Outputs at 0V
q
23.5
mA
V
S
=
5V, R
L
= 5k, C
L
= 30pF, Outputs at 0V
q
25.5
mA
Output Voltage Swing
V
S
=
2.375V, R
L
= 5k, C
L
= 30pF
q
4.0
4.6
V
P-P
V
S
=
5V, R
L
= 5k, C
L
= 30pF
q
9.3
9.8
V
P-P
V
OS
DC Offset Magnitude, V2 Outputs
V
S
=
2.375V, Input at AGND Voltage
q
3
15
mV
(Lowpass Response)
V
S
=
5V, Input at AGND Voltage
q
3
15
mV
DC AGND Reference Point
V
S
= Single 5V Supply
2.5
V
Center Frequency (f
O
) Error (Note 2)
LTC1562
V
S
=
5V, V2 Output Has R
L
= 5k, C
L
= 30pF
0.5
1.0
%
LTC1562A
V
S
=
5V, V2 Output Has R
L
= 5k, C
L
= 30pF
0.3
0.6
%
H
L
LP Passband Gain (V2 Output)
V
S
=
2.375V, f
IN
= 10kHz,
q
0
+ 0.05
+ 0.1
dB
V2 Output Has R
L
= 5k, C
L
= 30pF
H
B
BP Passband Gain (V1 Output)
V
S
=
2.375V, f
IN
= f
O
,
q
+ 0.2
+ 0.5
dB
V2 Output Has R
L
= 5k, C
L
= 30pF
(Note 1)
Total Supply Voltage (V
+
to V
) .............................. 11V
Maximum Input Voltage
at Any Pin ....................(V
0.3V)
V
(V
+
+ 0.3V)
Operating Temperature Range
LTC1562C ................................................ 0
C to 70
C
LTC1562I ............................................ 40
C to 85
C
Storage Temperature Range ................. 65
C to 150
C
Lead Temperature (Soldering, 10 sec) .................. 300
C
3
LTC1562
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Q Error
V
S
=
2.375V, LP Output Has R
L
= 5k, C
L
= 30pF
+ 3
%
Wideband Output Noise,
V
S
=
2.375V, BW = 200kHz, Input AC GND
24
V
RMS
Lowpass Response (V2 Output)
V
S
=
5V, BW = 200kHz, Input AC GND
24
V
RMS
Input-Referred Noise, Gain = 100
BW = 200kHz, f
O
= 100kHz, Q = 1, Input AC GND
4.5
V
RMS
THD
Total Harmonic Distortion,
f
IN
= 20kHz, 2.8V
P-P
, V1 and V2 Outputs Have
96
dB
Lowpass Response (V2 Output)
R
L
= 5k, C
L
= 30pF
f
IN
= 100kHz, 2.8V
P-P
, V1 and V2 Outputs Have
78
dB
R
L
= 5k, C
L
= 30pF
Shutdown Supply Current
SHDN Pin to V
+
1.5
15
A
SHDN Pin to V
+
, V
S
=
2.375V
1.0
A
Shutdown-Input Logic Threshold
2.5
V
Shutdown-Input Bias Current
SHDN Pin to 0V
10
20
A
Shutdown Delay
SHDN Pin Steps from 0V
to V
+
20
s
Shutdown Recovery Delay
SHDN Pin Steps from V
+
to 0V
100
s
Inverting Input Bias Current, Each Biquad
5
pA
ELECTRICAL CHARACTERISTICS
V
S
=
5V, outputs unloaded, T
A
= 25
C, SHDN pin to logic "low",
unless otherwise noted. AC specs are for a single 2nd order section, R
IN
= R2 = R
Q
=10k
0.1%, f
O
= 100kHz, unless noted.
The
q
denotes specifications that apply over the full operating
temperature range.
Note 1: Absolute Maximum Ratings are those values beyond which the life
of a device may be impaired.
Note 2: f
O
change from
5V to
2.375 supplies is 0.15% typical,
f
O
temperature coefficient, 40
C to 85
C, is 25ppm/
C typical.
TYPICAL PERFOR A CE CHARACTERISTICS
U
W
NOMINAL f
O
(kHz)
50
Q ERROR (%)
35
30
25
20
15
10
5
0
5
130
1562 G03
70
90
110
150
120
60
80
100
140
T
A
= 70
C
T
A
= 25
C
R
IN
= R
Q
Q = 10
Q = 5
Q = 2.5
Q = 1
Q Error vs Nominal f
O
(V
S
=
5V)
NOMINAL f
O
(kHz)
50
f
O
ERROR (%)
0
0.50
1.00
0.75
1.50
1.25
130
1562 G01
0.50
1.00
0.25
0.25
0.75
1.25
1.50
70
90
110
150
120
60
80
100
140
Q = 5
Q = 2.5
Q = 1
f
O
Error vs Nominal f
O
(V
S
=
5V)
NOMINAL f
O
(kHz)
50
f
O
ERROR (%)
0
0.50
1.00
0.75
1.50
1.25
130
1562 G02
0.50
1.00
0.25
0.25
0.75
1.25
1.50
70
90
110
150
120
60
80
100
140
Q = 5
Q = 2.5
Q = 1
f
O
Error vs Nominal f
O
(V
S
=
2.5V)
4
LTC1562
TYPICAL PERFOR A CE CHARACTERISTICS
U
W
Peak BP Gain vs Nominal f
O
(V
S
=
5V) (Figure 3, V1 Output)
Q Error vs Nominal f
O
(V
S
=
2.5V)
NOMINAL f
O
(kHz)
50
Q ERROR (%)
35
30
25
20
15
10
5
0
5
130
1562 G04
70
90
110
150
120
60
80
100
140
Q = 10
Q = 5
Q = 2.5
Q = 1
T
A
= 70
C
T
A
= 25
C
R
IN
= R
Q
NOMINAL f
O
(kHz)
50
0.5
PEAK BP GAIN (dB)
0
0.5
1.0
3.0
2.0
70
90 100
140
2.5
1.5
60
80
110 120 130
150
1562 G5
Q = 10
Q = 5
Q = 2.5
Q = 1
T
A
= 70
C
T
A
= 25
C
R
IN
= R
Q
NOMINAL f
O
(kHz)
50
0.5
PEAK BP GAIN (dB)
0
0.5
1.0
3.0
2.0
70
90 100
140
2.5
1.5
60
80
110 120 130
150
1562 G6
Q = 10
Q = 5
Q = 2.5
Q = 1
T
A
= 70
C
T
A
= 25
C
R
IN
= R
Q
Peak BP Gain vs Nominal f
O
(V
S
=
2.5V) (Figure 3, V1 Output)
Distortion vs External Load
Resistance (V
S
=
5V, 25
C)
(Figure 8)
EXTERNAL LOAD RESISTANCE (
)
10k
100
THD (AMPLITUDE BELOW FUNDAMENTAL) (dB)
80
70
60
50
40
30
5k
2k
1562 G09
20
10
0
90
1k
f
IN
= 50kHz
f
IN
= 20kHz
2nd ORDER LOWPASS
f
O
= 100kHz
Q = 0.7
OUTPUT LEVEL 1V
RMS
(2.83V
P-P
)
5V SUPPLIES
LP Noise vs Nominal f
O
(V
S
=
5V, 25
C) (Figure 3,
V2 Output) (R
IN
= R2)
NOMINAL f
O
(kHz)
60
10
BP NOISE (
V
RMS
)
15
25
30
35
60
45
80
100
110
1562 G08
20
50
55
40
70
90
120 130
140
Q = 5
Q = 2.5
Q = 1
PI
N
FU
N
CTIO
N
S
U
U
U
Power Supply Pins: The V
+
and V
pins should be
bypassed with 0.1
F capacitors to an adequate analog
ground or ground plane. These capacitors should be
connected as closely as possible to the supply pins. In the
20-lead SSOP package, the additional pins 4, 7, 14 and 17
are internally connected to V
(Pin 16) and should also be
tied to the same point as Pin 16 for best shielding. Low
noise linear supplies are recommended. Switching sup-
plies are not recommended as they will lower the filter
dynamic range.
Analog Ground (AGND): The AGND pin is the midpoint of
an internal resistive voltage divider, developing a potential
halfway between the V
+
and V
pins, with an equivalent
series resistance nominally 7k
. This serves as an inter-
nal ground reference. Filter performance will reflect the
quality of the analog signal ground and an analog ground
plane surrounding the package is recommended. The
analog ground plane should be connected to any digital
ground at a single point. For dual supply operation, the
AGND pin should be connected to the ground plane
BP Noise vs Nominal f
O
(V
S
=
5V, 25
C) (Figure 3,
V1 Output) (R
IN
= R
Q
)
NOMINAL f
O
(kHz)
60
10
NOISE (
V
RMS
)
15
25
30
35
60
45
80
100
110
1562 G07
20
50
55
40
70
90
120 130
140
Q = 5
Q = 2.5
Q = 1
5
LTC1562
PI
N
FU
N
CTIO
N
S
U
U
U
Shutdown (SHDN): When the SHDN input goes high or is
open-circuited, the LTC1562 enters a "zero-power" shut-
down state and only junction leakage currents flow. The
AGND pin and the amplifier outputs (see Figure 3) assume
a high impedance state and the amplifiers effectively
disappear from the circuit. (If an input signal is applied to
a complete filter circuit while the LTC1562 is in shutdown,
some signal will normally flow to the output through
passive components around the inactive op amps.)
A small pull-up current source at the SHDN input
defaults
the LTC1562 to the shutdown state if the SHDN pin is left
floating. Therefore, the user must connect the SHDN pin
to a logic "low" (0V for
5V supplies, V
for 5V total
supply) for normal operation of the LTC1562. (This con-
vention permits true "zero-power" shutdown since not
even the driving logic must deliver current while the part
is in shutdown.) With a single supply voltage, use V
for
logic "low"-- do not connect SHDN to the AGND pin.
(Figure 1). For single supply operation, the AGND pin
should be bypassed to the ground plane with at least a
0.1
F capacitor (at least 1
F for best AC performance)
(Figure 2).
Figure 1. Dual Supply Ground Plane Connection
(Including Substrate Pins 4, 7, 14, 17)
0.1
F
V
1562 F01
DIGITAL
GROUND PLANE
(IF ANY)
V
+
LTC1562
0.1
F
ANALOG
GROUND
PLANE
20
19
18
17
16
15
14
13
12
11
1
2
3
4
5
6
7
8
9
10
SINGLE-POINT
SYSTEM GROUND
1
F
1562 F01
DIGITAL
GROUND PLANE
(IF ANY)
V
+
LTC1562
V
+
/2
REFERENCE
0.1
F
ANALOG
GROUND
PLANE
20
19
18
17
16
15
14
13
12
11
1
2
3
4
5
6
7
8
9
10
SINGLE-POINT
SYSTEM GROUND
Figure 2. Single Supply Ground Plane Connection
(Including Substrate Pins 4, 7, 14, 17)
+
+
R2
R
Q
V
IN
INV
*R1 AND C ARE PRECISION
INTERNAL COMPONENTS
V2
V1
1/4 LTC1562
1562 F01
C
1
sR1C*
Z
IN
Z
IN
TYPE
R
C
RESPONSE
AT V1
BANDPASS
HIGHPASS
RESPONSE
AT V2
LOWPASS
BANDPASS
10k
R2
IN EACH CASE,
Q =
f
O
= (100kHz)
RQ
R2
( )
100kHz
f
O
( )
Figure 3. Equivalent Circuit of a Single 2nd Order Section
(Inside Dashed Line) Shown in Typical Connection. Form of Z
IN
Determines Response Types at the Two Outputs (See Table)
6
LTC1562
PI
N
FU
N
CTIO
N
S
U
U
U
INV A, INV B, INV C, INV D: Each of the INV pins is a virtual-
ground summing point for the corresponding 2nd order
section. For each section, external components Z
IN
, R2,
R
Q
connect to the INV pin as shown in Figure 3 and
described further in the Applications Information. Note
that the INV pins are sensitive internal nodes of the filter
and will readily receive any unintended signals that are
capacitively coupled into them. Capacitance to the INV
nodes will also affect the frequency response of the filter
sections. For these reasons, printed circuit connections to
the INV pins must be kept as short as possible, less than
one inch (2.5cm) total and surrounded by a ground plane.
V1 A, V1 B, V1 C, V1 D: Output Pins. Provide a bandpass,
highpass or other response depending on external cir-
cuitry (see Applications Information section). Each V1 pin
also connects to the R
Q
resistor of the corresponding 2nd
order filter section (see Figure 3 and Applications Informa-
tion). Each output is designed to drive a nominal net load
of 5k
and 30pF, which includes the loading due to the
external R
Q
. Distortion performance improves when the
outputs are loaded as lightly as possible. Some earlier
literature refers to these outputs as "BP" rather than V1.
V2 A, V2 B, V2 C, V2 D: Output Pins. Provide a lowpass,
bandpass or other response depending on external cir-
cuitry (see Applications Information section). Each V2 pin
also connects to the R2 resistor of the corresponding 2nd
order filter section (see Figure 3 and Applications Informa-
tion). Each output is designed to drive a nominal net load
of 5k
and 30pF, which includes the loading due to the
external R2. Distortion performance improves when the
outputs are loaded as lightly as possible. Some earlier
literature refers to these outputs as "LP" rather than V2.
BLOCK DIAGRA
W
Overall Block Diagram Showing Four 3-Terminal 2nd Order Sections
V
+
V
SHDN
1562 BD
2ND ORDER SECTIONS
R
R
INV
V1
V2
C
SHUTDOWN
SWITCH
SHUTDOWN
SWITCH
AGND
V
+
V
+
INV
V1
V2
INV
V1
V2
INV
V1
V2
C
C
C
A
B
D
C
+
+
+
7
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
Functional Description
The LTC1562 contains four matched, 2nd order, 3-termi-
nal universal continuous-time filter blocks, each with a
virtual-ground input node (INV) and two rail-to-rail out-
puts (V1, V2). In the most basic applications, one such
block and three external resistors provide 2nd order
lowpass and bandpass responses simultaneously (Figure
3, with a resistor for Z
IN
). The three external resistors set
standard 2nd order filter parameters f
O
, Q and gain. A
combination of internal precision components and exter-
nal resistor R2 sets the center frequency f
O
of each 2nd
order block. The LTC1562 is trimmed at manufacture so
that f
O
will be 100kHz
0.5% if the external resistor R2 is
exactly 10k.
However, lowpass/bandpass filtering is only one specific
application for the 2nd order building blocks in the LTC1562.
Highpass response results if the external impedance Z
IN
in
Figure 3 becomes a capacitor C
IN
(whose value sets only
gain, not critical frequencies) as described below.
Responses with zeroes are available through other con-
nections (see Notches and Elliptic Responses). Moreover,
the virtual-ground input gives each 2nd order section the
built-in capability for analog operations such as gain
(preamplification), summing and weighting of multiple
inputs, handling input voltages beyond the power supplies
or accepting current or charge signals directly. These
Operational Filter
TM
frequency-selective building blocks
are nearly as versatile as operational amplifiers.
The user who is not copying exactly one of the Typical
Applications schematics shown later in this data sheet is
urged to read carefully the next few sections through at
least Signal Swings, for orientation about the LTC1562,
before attempting to design custom application circuits.
Also available free from LTC, and recommended for de-
signing custom filters, is the general-purpose analog filter
design software FilterCAD
TM
for Windows
. This software
includes tools for finding the necessary f
0
, Q and gain
parameters to meet target filter specifications such as
frequency response.
Setting f
O
and Q
Each of the four 2nd order sections in the LTC1562 can be
programmed for a standard filter function (lowpass,
bandpass or highpass) when configured as in Figure 3
with a resistor or capacitor for Z
IN
. These transfer func-
tions all have the same denominator, a complex pole pair
with center frequency
O
= 2
f
O
and quality parameter Q.
(The numerators depend on the response type as de-
scribed below.) External resistors R2 and R
Q
set f
O
and Q
as follows:
f
C R R
k
R
kHz
O
=
=
(
)
1
2
1 2
10
2
100
( )
Q
R
R R
R
k
R
R
R
kHz
f
Q
Q
Q
O
=
=
=
( )
(
)
1 2
10
2
2
100
R1 = 10k and C = 159pF are internal to the LTC1562 while
R2 and R
Q
are external.
A typical design procedure proceeds from the desired f
O
and Q as follows, using finite-tolerance fixed resistors.
First find the ideal R2 value for the desired f
O
:
R Ideal
kHz
f
k
O
2
100
10
2
( )
=
( )
Then select a practical R2 value from the available finite-
tolerance resistors. Use the actual R2 value to find the
desired R
Q
, which also will be approximated with finite
tolerance:
R
Q
k
R
Q
=
(
)
10
2
The f
O
range is approximately 10kHz to 150kHz, limited
mainly by the magnitudes of the external resistors
required. As shown above, R2 varies with the inverse
square of f
O
. This relationship desensitizes f
O
to R2's
Operational Filter and FilterCAD are trademarks of Linear Technology Corporation.
Windows is a registered trademark of Microsoft Corporation.
8
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
tolerance (by a factor of 2 incrementally), but it also
implies that R2 has a wider range than f
O
. (R
Q
and R
IN
also
tend to scale with R2.) At high f
O
these resistors fall below
5k, heavily loading the outputs of the LTC1562 and leading
to increased THD and other effects. At the other extreme,
a lower f
O
limit of 10kHz reflects an arbitrary upper
resistor limit of 1M
. The LTC1562's MOS input circuitry
can accommodate higher resistor values than this, but
junction leakage current from the input protection cir-
cuitry may cause DC errors.
The 2nd order transfer functions H
LP
(s), H
BP
(s) and
H
HP
(s) (below) are all inverting so that, for example, at DC
the lowpass gain is H
L
. If two such sections are cas-
caded, these phase inversions cancel. Thus, the filter in the
application schematic on the first page of this data sheet
is a dual DC preserving, noninverting, rail-to-rail lowpass
filter, approximating two "straight wires with frequency
selectivity."
Figure 4 shows further details of 2nd order lowpass,
bandpass and highpass responses. Configurations to
obtain these responses appear in the next three sections.
Basic Lowpass
When Z
IN
of Figure 3 is a resistor of value R
IN
, a standard
2nd order lowpass transfer function results from V
IN
to V2
(Figure 5):
V s
V s
H
s
H
s
Q s
IN
LP
L
O
O
O
2
2
2
2
( )
( )
( )
/
=
=
+
( )
+
The DC gain magnitude is H
L
= R2/R
IN
. (Note that the
transfer function includes a sign inversion.) Parameters
O
(= 2
f
O
) and Q are set by R2 and R
Q
as above. For a 2nd
order lowpass response the gain magnitude becomes QH
L
INV
V1
2nd ORDER
1/4 LTC1562
V2
1562 F05
R2
R
Q
R
IN
V
IN
V
OUT
f
L
GAIN (V/
V)
0.707 H
B
H
B
f
O
f (LOG SCALE)
BANDPASS RESPONSE
f
H
GAIN (V/
V)
0.707 H
L
H
P
H
L
H
H
f
P
f (LOG SCALE)
LOWPASS RESPONSE
f
C
f
C
GAIN (V/
V)
0.707 H
H
H
P
f
P
f (LOG SCALE)
HIGHPASS RESPONSE
Q
f
f
f
f
f f
f
f
Q
Q
f
f
Q
Q
O
H
L
O
L H
L
O
H
O
=
=
=
+
+
=
+
+
;
1
2
1
2
1
1
2
1
2
1
2
2
f
f
Q
Q
f
f
Q
H
H
Q
Q
C
O
P
O
P
L
=
+
+
=
=
1
1
2
1
1
2
1
1
1
2
1
1
1
1
4
2
2
2
2
2
f
f
Q
Q
f
f
Q
H
H
Q
Q
C
O
P
O
P
H
=
+
+
=
=
1
1
2
1
1
2
1
1
1
2
1
1
1
1
4
2
2
2
1
2
1
2
Figure 4. Characteristics of Standard 2nd Order Filter Responses
Figure 5. Basic Lowpass Configuration
9
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
Parameters
O
= 2
f
O
and Q are set by R2 and R
Q
as
above. The highpass gain parameter is H
H
= C
IN
/159pF.
For a 2nd order highpass response the gain magnitude at
frequency f
O
is QH
H
, and approaches H
H
at high frequen-
cies (f >> f
O
). For Q > 0.707, a gain peak occurs at a
frequency above f
O
as shown in Figure 4. The transfer
function includes a sign inversion.
at frequency f
O
, and for Q > 0.707, a gain peak occurs at
a frequency below f
O
, as shown in Figure 4.
Basic Bandpass
There are two different ways to obtain a bandpass function
in Figure 3, both of which give the following transfer
function form:
H
s
H
Q s
s
Q s
BP
B
O
O
O
( )
/
/
=
( )
+
( )
+
2
2
O
= 2
f
O
and Q are set by R2 and R
Q
as described previ-
ously in Setting f
O
and Q. When Z
IN
is a resistor of value
R
IN
, a bandpass response results at the V1 output (Figure
6a) with a gain parameter H
B
= R
Q
/R
IN
. Alternatively, a
capacitor of value C
IN
gives a bandpass response at the V2
output (Figure 6b), with the same H
BP
(s) expression, and
the gain parameter now H
B
= (R
Q
/10k
)(C
IN
/159pF). This
transfer function has a gain magnitude of H
B
(its peak value)
when the frequency equals f
O
and has a phase shift of 180
at that frequency. Q measures the sharpness of the peak
(the ratio of f
O
to 3dB bandwidth) in a 2nd order bandpass
function, as illustrated in Figure 4.
INV
V1
2nd ORDER
1/4 LTC1562
(b) Capacitive Input
(a) Resistive Input
V2
1562 F06
R2
R
Q
C
IN
V
IN
V
OUT
INV
V1
2nd ORDER
1/4 LTC1562
V2
R2
R
Q
R
IN
V
IN
V
OUT
Figure 6. Basic Bandpass Configurations
Basic Highpass
When Z
IN
of Figure 3 is a capacitor of value C
IN
, a highpass
response appears at the V1 output (Figure 7).
V s
V s
H
s
H s
s
Q s
IN
HP
H
O
O
1
2
2
2
( )
( )
( )
/
=
=
+
( )
+
INV
V1
2nd ORDER
1/4 LTC1562
V2
1562 F07
R2
R
Q
C
IN
V
IN
V
OUT
Figure 7. Basic Highpass Configuration
Signal Swings
The V1 and V2 outputs are capable of swinging to within
roughly 100mV of each power supply rail. As with any
analog filter, the signal swings in each 2nd order section
must be scaled so that no output overloads (saturates),
even if it is not used as a signal output. (Filter literature
often calls this the "dynamics" issue.) When an unused
output has a larger swing than the output of interest, the
section's gain or input amplitude must be scaled down to
avoid overdriving the unused output. The LTC1562 can
still be used with high performance in such situations as
long as this constraint is followed.
For an LTC1562 section as in Figure 3, the magnitudes of
the two outputs V2 and V1, at a frequency
= 2
f, have
the ratio,
|
(
)|
|
(
)|
(
)
V j
V j
kHz
f
2
1
100
=
regardless of the details of Z
IN
. Therefore, an input fre-
quency above or below 100kHz produces larger output
amplitude at V1 or V2, respectively. This relationship can
guide the choice of filter design for maximum dynamic
range in situations (such as bandpass responses) where
there is more than one way to achieve the desired fre-
quency response with an LTC1562 section.
10
LTC1562
Because 2nd order sections with Q
1 have response
peaks near f
O
, the gain ratio above implies some rules of
thumb:
f
O
< 100kHz
V2 tends to have the larger swing
f
O
> 100kHz
V1 tends to have the larger swing.
The following situations are convenient because the
relative swing issue does not arise. The unused output's
swing is naturally the smaller of the two in these cases:
Lowpass response (resistor input, V2 output, Figure 5)
with f
O
< 100kHz
Bandpass response (capacitor input, V2 output, Figure
6b) with f
O
< 100kHz
Bandpass response (resistor input, V1 output, Figure
6a) with f
O
> 100kHz
Highpass response (capacitor input, V1 output, Figure
7) with f
O
> 100kHz
The LTC1562-2, a higher frequency derivative of the
LTC1562, has a design center f
O
of 200kHz compared to
100kHz in the LTC1562. The rules summarized above
apply to the LTC1562-2 but with 200kHz replacing the
100kHz limits. Thus, an LTC1562-2 lowpass filter section
with f
O
below 200kHz automatically satisfies the desirable
condition of the unused output carrying the smaller signal
swing.
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
level inputs require further dynamic range, reducing the
value of Z
IN
boosts the signal gain while reducing the input
referred noise. This feature can increase the SNR for low
level signals. Varying or switching Z
IN
is also an efficient
way to effect automatic gain control (AGC). From a system
viewpoint, this technique boosts the ratio of maximum
signal to minimum noise, for a typical 2nd order lowpass
response (Q = 1, f
O
= 100kHz), to 118dB.
Input Voltages Beyond the Power Supplies
Properly used, the LTC1562 can accommodate input
voltage excursions well beyond its supply voltage. This
requires care in design but can be useful, for example,
when large out-of-band interference is to be removed from
a smaller desired signal. The flexibility for different input
voltages arises because the INV inputs are at virtual
ground potential, like the inverting input of an op amp with
negative feedback. The LTC1562 fundamentally responds
to input
current and the external voltage V
IN
appears only
across the external impedance Z
IN
in Figure 3.
To accept beyond-the-supply input voltages, it is impor-
tant to keep the LTC1562 powered on, not in shutdown
mode, and to avoid saturating the V1 or V2 output of the
2nd order section that receives the input. If any of these
conditions is violated, the INV input will depart from a
virtual ground, leading to an overload condition whose
recovery timing depends on circuit details. In the event
that this overload drives the INV input beyond the supply
voltages, the LTC1562 could be damaged.
The most subtle part of preventing overload is to consider
the possible input signals or spectra and take care that
none of them can drive either V1 or V2 to the supply limits.
Note that neither output can be allowed to saturate, even
if it is not used as the signal output. If necessary the
passband gain can be reduced (by increasing the imped-
ance of Z
IN
in Figure 3) to reduce output swings.
The final issue to be addressed with beyond-the-supply
inputs is current and voltage limits. Current entering the
virtual ground INV input flows eventually through the
output circuitry that drives V1 and V2. The input current
magnitude (
V
IN
/
Z
IN
in Figure 3) should be limited by
design to less than 1mA for good distortion performance.
On the other hand, the input voltage V
IN
appears across the
Low Level or Wide Range Input Signals
The LTC1562 contains a built-in capability for low noise
amplification of low level signals. The Z
IN
impedance in
each 2nd order section controls the block's gain. When set
for unity passband gain, a 2nd order section can deliver an
output signal more than 100dB above the noise level. If low
Figure 8. 100kHz, Q = 0.7 Lowpass Circuit for
Distortion vs Loading Test
INV
V1
2nd ORDER
1/4 LTC1562
V2
1562 F08
R2
10k
C
L
30pF
R
L
(EXTERNAL
LOAD RESISTANCE)
R
Q
6.98k
R
IN
10k
V
IN
V
OUT
11
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
external component Z
IN
, usually a resistor or capacitor.
This component must of course be rated to sustain the
magnitude of voltage imposed on it.
Lowpass "T" Input Circuit
The virtual ground INV input in the Operational Filter block
provides a means for adding an "extra" lowpass pole to
any resistor-input application (such as the basic lowpass,
Figure 5, or bandpass, Figure 6a). The resistor that would
otherwise form Z
IN
is split into two parts and a capacitor
to ground added, forming an R-C-R "T" network (Figure
9). This adds an extra, independent real pole at a fre-
quency:
f
R C
P
P T
=
1
2
where C
T
is the new external capacitor and R
P
is the
parallel combination of the two input resistors R
INA
and
R
INB
. This pair of resistors must normally have a pre-
scribed series total value R
IN
to set the filter's gain as
described above. The parallel value R
P
can however be set
arbitrarily (to R
IN
/4 or less) which allows choosing a
convenient standard capacitor value for C
T
and fine tuning
the new pole with R
P
.
INV
V1
2nd ORDER
1/4 LTC1562
V2
1562 F09
R2
R
Q
R
INB
R
INA
C
T
V
IN
Figure 9. Lowpass "T" Input Circuit
The procedure therefore is to begin with the target extra
pole frequency f
P
. Determine the series value R
IN
from the
gain requirement. Select a capacitor value C
T
such that R
P
= 1/(2
f
P
C
T
) is no greater than R
IN
/4, and then choose
R
INA
and R
INB
that will simultaneously have the parallel
value R
P
and the series value R
IN
. Such R
INA
and R
INB
can
be found directly from the expression:
1
2
1
2
4
2
R
R
R R
IN
IN
IN P
(
)
A practical limitation of this technique is that the C
T
capaci-
tor values that tend to be required (hundreds or thousands
of pF) can destabilize the op amp in Figure 3 if R
INB
is too
small, leading to AC errors such as Q enhancement. For this
reason, when R
INA
and R
IN
B are unequal, preferably the
larger of the two should be placed in the R
INB
position.
Highpass "T" Input Circuit
A method similar to the preceding technique adds an
"extra" highpass pole to any capacitor-input application
(such as the bandpass of Figure 6b or the highpass of
Figure 7). This method splits the input capacitance C
IN
into
two series parts C
INA
and C
INB
, with a resistor R
T
to ground
between them (Figure 10). This adds an extra 1st order
highpass corner with a zero at DC and a pole at the
frequency:
f
R C
P
T P
=
1
2
where C
P
= C
INA
+ C
INB
is the parallel combination of the
two capacitors. At the same time, the total series capaci-
tance C
IN
will control the filter's gain parameter (H
H
in
Basic Highpass). For a given series value C
IN
, the parallel
value C
P
can still be set arbitrarily (to 4C
IN
or greater).
Figure 10. Highpass "T" Input Circuit
INV
V1
2nd ORDER
1/4 LTC1562
V2
1562 F10
R2
R
Q
C
INB
R
T
V
IN
C
INA
The procedure then is to begin with the target corner (pole)
frequency f
P
. Determine the series value C
IN
from the gain
requirement (for example, C
IN
= H
H
(159pF) for a highpass).
Select a resistor value R
T
such that C
P
= 1/(2
R
T
f
P
) is at
least 4C
IN
, and select C
INA
and C
INB
that will simultaneously
have the parallel value C
P
and the series value C
IN
. Such
C
INA
and C
INB
can be found directly from the expression:
1
2
1
2
4
2
C
C
C C
P
P
IN P
(
)
12
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
3dB frequencies f
L
and f
H
are widely separated from this
peak.
The LTC1562's f
O
is trimmed in production to give an
accurate 180
phase shift in the configuration of Figure
6a with resistor values setting f
0
= 100kHz and Q = 1.
Table 1 below shows typical differences between f
O
values measured via the bandpass 180
criterion and f
O
values measured using the two other methods listed
above (Figure 6a, R
IN
= R
Q
).
Table 1
f
O
Q = 1
Q = 1
Q = 5
Q = 5
(BP 180
)
BP-PEAK f
O
L
H
f
O
BP-PEAK f
O
L
H
f
O
60kHz
+ 0.3%
+ 0.3%
+ 0.05%
+ 0.05%
100kHz
+ 0.6%
+ 0.6%
+ 0.1%
+ 0.1%
140kHz
+ 0.8%
+ 0.8%
+ 0.15%
+ 0.15%
LTC1562 Demo Board
The LTC1562 demo board is assembled with an LTC1562
or LTC1562A in a 20-pin SSOP package and power supply
decoupling capacitors. Jumpers on the board configure
the LTC1562 for dual or single supply operation and power
shutdown. Pads for surface mount resistors and capaci-
tors are provided to build application-specific filters. Also
provided are terminals for inputs, outputs and power
supplies.
This procedure can be iterated, adjusting the value of R
T
,
to find convenient values for C
INA
and C
INB
since resistor
values are generally available in finer increments than
capacitor values.
Different "f
O
" Measures
Standard 2nd order filter algebra, as in Figure 4 and the
various transfer-function expressions in this data sheet,
uses a center frequency parameter f
O
(or
O
, which is
2
f
O
). f
O
can also be measured in practical ways, includ-
ing:
The frequency where a bandpass response has 180
phase shift
The frequency where a bandpass response has peak
gain
The geometric mean of the 3.01dB gain frequencies in
a bandpass (
L
H
in Figure 4)
An ideal mathematical 2nd order response yields exactly
the same frequency by these three measures. However,
real 2nd order filters with finite-bandwidth circuitry show
small differences between the practical f
O
measures,
which may be important in critical applications. The issue
is chiefly of concern in high-Q bandpass applications
where, as the data below illustrate, the different f
0
mea-
surements tend to converge anyway for the LTC1562. At
low Q the bandpass peak is not sharply defined and the
13
LTC1562
TYPICAL APPLICATIO
N
S
U
(Basic)
Quad 3rd Order Butterworth Lowpass Filter, Gain = 1
Quad 3rd Order
Butterworth
f
3dB
f
3dB
f
3dB
f
3dB
f
3dB
f
3dB
f
3dB
Lowpass Filters
20kHz
40kHz
60kHz
80kHz
100kHz
120kHz
140kHz
C
IN
220pF
1000pF
1000pF
1000pF
1000pF
1000pF
1000pF
R
INA
44.2k
4.32k
3.16k
2.43k
1.96k
1.87k
1.69k
R
INB
205k
57.6k
24.3k
13.0k
8.06k
5.11k
3.4k
R
Q
249k
61.9k
27.4k
15.4k
10.0k
6.98k
5.11k
R2
249k
61.9k
27.4k
15.4k
10.0k
6.98k
5.11k
All four sections have identical R
INA
, R
INB
and C
IN
values. All resistor values are
1%
FREQUENCY (Hz)
10k
GAIN (dB)
10
0
10
20
30
40
50
60
100k
1M
1562 TA05b
f
3dB
= 100kHz
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN1B
R
IN1A
R
IN3A
V
IN1
V
IN3
C
IN1
V
IN2
1562 TA05a
V
OUT2
V
OUT3
V
OUT4
V
OUT1
R
IN3B
5V
5V
R
Q1
R21
R23
0.1
F
0.1
F
R
Q3
R24
R
Q4
R
Q2
R22
C
IN2
C
IN3
R
IN2B
R
IN2A
V
IN4
C
IN4
R
IN4B
R
IN4A
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
Amplitude Response
14
LTC1562
TYPICAL APPLICATIO
N
S
U
(Basic)
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
R
IN4
R
IN1
V
IN2
V
IN1
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
V
OUT1
1562 TA03a
V
OUT2
R
IN3
5V
5V
R
Q1
R21
R23
0.1
F
0.1
F
R
Q3
R24
R
Q4
R
Q2
R22
Dual 4th Order Lowpass Filters
Amplitude Response
FREQUENCY (Hz)
10k
GAIN (dB)
10
0
10
20
30
40
50
60
70
80
100k
1M
1562 TA03b
BUTTERWORTH
f
3dB
= 100kHz
10k
R21, R23, R
IN1
, R
IN3
=
Quick Design Formulas for Some Popular Response Types:
2
100kHz
C
Butterworth
(Maximally Flat Passband)
for f
C
10kHz to 140kHz
14.24k
2
100kHz
C
Chebyshev
(Equiripple Passband)
for f
C
20kHz to 120kHz
3.951k
2
100kHz
C
Bessel
(Good Transient Response)
for f
C
10kHz to 70kHz
5.412k
R
Q1
, R
Q3
=
100kHz
C
7.26k
100kHz
C
5.066k
100kHz
C
10k
R22, R24, R
IN2
, R
IN4
=
2
100kHz
C
7.097k
2
100kHz
C
4.966k
2
100kHz
C
13.07k
R
Q2
, R
Q4
=
Notes: f
C
is the cutoff frequency: For Butterworth and Bessel, response is 3dB down at f
C
. For Chebyshev filters with
0.1dB passband ripple up to 0.95 f
C
, use LTC1562 "A" grade.
Example: Butterworth response, f
C
= 50kHz. from the formulas above, R21 = R23 = R
IN1
= R
IN3
= 10k(100kHz/50kHz)
2
= 40k. R
Q1
= R
Q3
= 5.412k(100kHz/50kHz) = 10.82k. R22 = R24 = R
IN2
= R
IN4
= 10k(100kHz/50kHz)
2
= 40k.
R
Q2
= R
Q4
= 13.07k(100kHz/50kHz) = 26.14k. Use nearest 1% values.
100kHz
C
17.53k
100kHz
C
3.679k
100kHz
C
1562 TA03 TABLE
15
LTC1562
TYPICAL APPLICATIO
N
S
U
(Basic)
8th Order Lowpass Filters
Amplitude Response
R21 = R
IN1
= 10k
Quick Design Formulas for Some Popular Response Types:
2
100kHz
C
Butterworth
(Maximally Flat Passband)
for f
C
10kHz to 140kHz
R21 = 7.51k
, R
IN1
= 2.2R21*
, R
IN4
=
2
100kHz
C
Chebyshev
(Equiripple Passband)
for f
C
20kHz to 120kHz
Bessel
(Good Transient Response)
for f
C
10kHz to 70kHz
R
Q1
= 6.01k
100kHz
C
R
Q1
= 119.3k
100kHz
C
100kHz
C
+ 560kHz
100kHz
C
+ 530kHz
R24*
2.2
100kHz
C
+ 2440kHz
R22 = R
IN2
= 10k
2
100kHz
C
R22 = R
IN2
= 14.99k
2
100kHz
C
R
Q2
= 9k
Notes: f
C
is the cutoff frequency: For Butterworth and Bessel, response is 3dB down at f
C
. For Chebyshev filters with
0.1dB passband ripple up to 0.95 f
C
, use LTC1562 "A" grade. *The resistor values marked with an asterisk (*) in the
Chebyshev formulas (R21 and R24) should be rounded to the nearest standard finite-tolerance value
before
computing
the values dependent on them (R
IN1
and R
IN4
respectively).
Example: Chebyshev response, f
C
= 100kHz. The formulas above give R21 = 7.51k, nearest standard 1% value 7.50k.
Using this 1% value gives R
IN1
= 16.5k, already a standard 1% value. R
Q1
= 18.075k, nearest 1% value 18.2k.
R22 = R
IN2
= 14.99k, nearest 1% value 15k. R
Q2
= 11.02k, nearest 1% value 11k. R23 = R
IN3
= 7.15k, already a
standard 1% value. R
Q3
= 18.75k, nearest 1% value 18.7k. R24 = 26.7k, already a standard 1% value. This gives
R
IN4
= 12.14k, nearest 1% value 12.1k. R
Q4
= 8.75k, nearest 1% value 8.66k.
100kHz
C
R
Q2
= 279.9k
100kHz
C
R23 = R
IN3
= 10k
2
100kHz
C
R23 = R
IN3
= 7.15k
2
100kHz
C
R
Q3
= 5.1k
100kHz
C
R
Q3
= 118.1k
100kHz
C
R24 = R
IN4
= 10k
2
100kHz
C
R24 = 26.7k
2
100kHz
C
R
Q4
= 25.63k
100kHz
C
R21 = R
IN1
= 2.61k
2
100kHz
C
R
Q1
= 3.63k
100kHz
C
R22 = R
IN2
= 2.07k
2
100kHz
C
R
Q2
= 5.58k
100kHz
C
R23 = R
IN3
= 2.96k
2
100kHz
C
R
Q3
= 3.05k
100kHz
C
R24 = R
IN4
= 3.14k
2
100kHz
C
R
Q4
= 2.84k
100kHz
C
R
Q4
= 8.75k
100kHz
C
1562 TA04 TABLE
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
R
IN4
R
IN1
V
IN
V
OUT
1562 TA04a
R
IN3
5V
5V
R
Q1
R21
R23
0.1
F
0.1
F
R
Q3
R24
R
Q4
R
Q2
R22
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
FREQUENCY (Hz)
GAIN (dB)
10
0
10
20
30
40
50
60
70
80
90
10k
100k
500k
1562 TA04b
CHEBYSHEV
f
C
= 100kHz
16
LTC1562
(Basic)
TYPICAL APPLICATIO
N
S
U
Amplitude Response
8th Order Bandpass Filter, Single 5V Supply,
3dB Bandwidth =
Center Frequency
10
R21 = R23 = 10.6k
Quick Design Formulas for Center Frequency f
C
(Recommended Range 40kHz to 140kHz):
2
100kHz
C
R
Q1
= R
Q3
= 164.6k
100kHz
C
100kHz
C
+ 319kHz
R
Q2
= R
Q4
= 143.2k
R
IN2
= R
IN4
=
100kHz
C
+ 294kHz
100kHz
C
+ 286kHz
R22 = R24 = 9.7k
100kHz
C
100kHz
C
C
IN1
= C
IN3
= 159pF
2
10k
R
Q1
R22R
Q1
C
IN1
(10k)(10.6pF)
Notes: R
Q1
, R22 and C
IN1
should be rounded to the nearest standard finite-tolerance value
before using these
values in the later formulas.
Example: Center frequency f
C
of 80kHz. The formulas give R21 = R23 = 16.56k, nearest standard 1% value 16.5k.
R
Q1
= R
Q3
= 51.56k, nearest 1% value 51.1k. R22 = R24 = 15.15k, nearest 1% value 15k. R
Q2
= R
Q4
= 47.86k,
nearest 1% value 47.5k. C
IN1
= C
IN2
= 31.11pF using 51.1k for R
Q1
, nearest standard 5% capacitor value 33pF.
This and the 1% value R22 = 15k also go into the calculation for R
IN2
= R
IN4
= 65.20k, nearest 1% value 64.9k.
1562 TA07 TABLE
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
R
IN4
V
IN
V
OUT
1562 TA07a
5V
R
Q1
R21
R23
0.1
F
1
F
R
Q3
C
IN3
C
IN1
R24
R
Q4
R
Q2
R22
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
FREQUENCY (kHz)
40
GAIN (dB)
30
10
10
104
1562 TA07b
50
70
40
20
0
60
80
90
56
72
88
48
112
64
80
96
120
f
CENTER
= 80kHz
17
LTC1562
(Basic)
TYPICAL APPLICATIO
N
S
U
Amplitude Response
8th Order Bandpass Filter, Single 5V Supply,
1dB Bandwidth =
Center Frequency
10
FREQUENCY (kHz)
60
GAIN (dB)
30
10
10
124
1562 TA06b
50
70
40
20
0
60
80
90
76
92
108
68
132
84
100
116
140
f
CENTER
= 100kHz
R21 = R23 = 11.7k
Quick Design Formulas for a Center Frequency f
C
(Recommended Range 50kHz to 120kHz):
2
2
100kHz
C
R
IN1
= R
IN3
=
R21
2.56
R22 = R24 = 8.66k
100kHz
C
C
+ 1736kHz
100kHz
R
IN2
= R
IN4
=
R
Q2
14.36
C
+ 634kHz
100kHz
Notes: R21 and R
Q2
should be rounded to the nearest standard finite-tolerance value
before using these values in
the later formulas. For f
C
< 100kHz, the maximum peak-to-peak passband input level is (f
C
/100kHz)5V. Use
LTC1562A for minimum variation of passband gain.
Example: Center frequency f
C
of 100kHz. The formulas give R21 = R23 = 11.7k, nearest standard 1% value 11.5k.
This value gives R
IN1
= R
IN3
= 82.46k, nearest 1% value 82.5k. R
Q1
= R
Q3
= 65.5k, nearest 1% value 64.9k.
R22 = R24 = 8.66k, already a standard 1% value. This gives R
IN2
= R
IN4
= 32.4k (again already a standard 1% value).
R
Q2
= R
Q4
= 63.45k, nearest 1% value 63.4k. If LTC1562A is used, resistor tolerances tighter than 1% will further
improve the passband gain accuracy.
R
Q1
= R
Q3
= 215.5k
100kHz
C
100kHz
C
1562 TA06 TABLE
100kHz
C
+ 229kHz
R
Q2
= R
Q4
= 286.2k
100kHz
C
+ 351kHz
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
R
IN4
R
IN1
V
IN
V
OUT
1562 TA06a
R
IN3
5V
R
Q1
R21
R23
0.1
F
1
F
R
Q3
R24
R
Q4
R
Q2
R22
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
18
LTC1562
(Basic)
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
R
IN4
R
IN1
V
IN
V
OUT
1562 TA08a
R
IN3
V
V
+
R
Q1
R21
R23
0.1
F
0.1
F
R
Q3
R24
R
Q4
R
Q2
R22
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
FREQUENCY (kHz)
40
GAIN (dB)
10
10
30
120
1562 TA08b
30
50
20
0
20
40
60
70
60
80
100
140
160
180
f
CENTER
= 100kHz
Amplitude Response
8th Order Bandpass Filter
3dB BW =
f
CENTER
, Gain = 10
f
CENTER
f
CENTER
f
CENTER
f
CENTER
f
CENTER
f
CENTER
f
CENTER
10
80kHz
90kHz
100kHz
110kHz
120kHz
130kHz
140kHz
Side B
R
IN1
4.64k
5.23k
6.34k
5.11k
5.11k
5.49k
5.62k
R
Q1
46.4k
52.3k
42.2k
38.3k
34.8k
32.4k
30.1k
R21
12.4k
15.4k
10.0k
8.25k
6.98k
5.9k
5.11k
Sides A, C, D
R
IN2
, R
IN3
, R
IN4
46.4k
52.3k
42.2k
38.3k
34.8k
32.4k
30.1k
R
Q2
, R
Q3
, R
Q4
46.4k
52.3k
42.2k
38.3k
34.8k
32.4k
30.1k
R22, R23, R24
12.4k
15.4k
10.0k
8.25k
6.98k
5.90k
5.11k
All resistor values are
1%
TYPICAL APPLICATIO
N
S
U
8th Order Bandpass (High Frequency) Filter
3dB Bandwidth =
Center Frequency
, Gain = 10
10
19
LTC1562
(Basic)
TYPICAL APPLICATIO
N
S
U
Amplitude Response
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
,
5.23k
R
IN4
,
3.4k
C
IN1
150pF
V
IN
V
OUT
1562 TA11a
R
IN3
, 8.06k
5V
5V
R
Q1
, 30.1k
R21, 110k
R23, 5.23k
0.1
F
0.1
F
R
Q3
, 14k
R24, 5.23k
R
Q4
, 3.74k
R
Q2
, 5.11k
R22, 5.23k
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
ALL RESISTORS = 1% METAL FILM
2nd Order 30kHz Highpass Cascaded with 6th Order 138kHz Lowpass
8th Order Wideband Bandpass Filter
f
CENTER
= 50kHz, 3dB BW 40kHz to 60kHz
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
C
IN1
22pF
V
IN
V
OUT
1562 TA09a
V
+
V
R
Q1
59k
R21 56.2k
R
IN2
69.8k
R23 63.4k
0.1
F
1
F
R
Q3
82.5k
R24 28.7k
R
Q4
100k
R
Q2
48.7k
R22 34.8k
C
IN4
47pF
C
IN3
27pF
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
FREQUENCY (kHz)
60
30
40
50
10
0
10
20
1562 TA09b
GAIN (dB)
20
100
Amplitude Response
FREQUENCY (kHz)
10
GAIN (dB)
20
10
0
10
20
30
40
50
60
70
80
100
400
1562 TA11b
8th Order Highpass 0.05dB Ripple Chebyshev Filter f
CUTOFF
= 30kHz
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
C
IN1
150pF
C
IN
1562 TA10a
V
OUT
5V
5V
R
Q1
, 10.2k
R21, 35.7k
R23, 107k
0.1
F
0.1
F
R
Q3
, 54.9k
R24, 127k
R
Q4
, 98.9k
R
Q2
, 22.1k
R22, 66.5k
C
IN3
150pF
C
IN4
150pF
C
IN2
150pF
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
TOTAL OUTPUT NOISE = 40
V
RMS
Amplitude Response
FREQUENCY (Hz)
1k
GAIN (dB)
10
0
10
20
30
40
50
60
70
80
90
10k
100k
1M
1562 TA10b
20
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
Notches and Elliptic Responses
The basic (essentially all-pole) LTC1562 circuit tech-
niques described so far will serve many applications.
However, the sharpest-cutoff lowpass, highpass and
bandpass filters include notches (imaginary zero pairs) in
the stopbands. A notch, or band-reject, filter has zero gain
at a frequency f
N
. Notches are also occasionally used by
themselves to reject a narrow band of frequencies. A
number of circuit methods will give notch responses from
an Operational Filter block. Each method exhibits an input-
output transfer function that is a standard 2nd order band-
reject response:
H
s
H s
s
Q s
BR
N
N
O
O
( )
/
=
+
+
( )
+
2
2
2
2
with parameters
N
= 2
f
N
and H
N
set by component
values as described below. (
0
= 2
f
0
and Q are set for the
Operational Filter block by its R2 and R
Q
resistors as
described earlier in Setting f
0
and Q). Characteristically,
the gain magnitude |H
BR
(j2
f)| has the value H
N
(f
N
2
/f
0
2
) at
DC (f = 0) and H
N
at high frequencies (f >> f
N
), so in
addition to the notch, the gain changes by a factor:
High Frequency Gain
DC Gain
O
N
=
2
2
The common principle in the following circuit methods is
to add a signal to a filtered replica of itself having equal gain
and 180
phase difference at the desired notch frequency
f
N
. The two signals then cancel out at frequency f
N
. The
notch depth (the completeness of cancellation) will be
infinite to the extent that the two paths have matching
gains. Three practical circuit methods are presented here,
with different features and advantages.
Examples and design procedures for practical filters using
these techniques appear in a series of articles
1
attached to
this data sheet on the Linear Technology web site
(www.linear-tech.com). Also available free is the analog
filter design software, FilterCAD for Windows, recom-
mended for designing filters not shown in the Typical
Applications schematics in this data sheet.
Elementary Feedforward Notches
A "textbook" method to get a 180
phase difference at
frequency f
N
for a notch is to dedicate a bandpass 2nd
order section (described earlier under Basic Bandpass),
which gives 180
phase shift at the section's center
frequency f
O
(Figure 11, with C
IN1
= 0), so that f
N
= f
O
. The
bandpass section of Figure 6a, at its center frequency f
O
,
has a phase shift of 180
and a gain magnitude of H
B
=
R
Q
/R
IN
. A notch results in Figure 11 if the paths summed
into virtual ground have the same gains at the 180
frequency (then I
O
= 0). This requires a constraint on the
resistor values:
R
R
R
R
IN
FF
Q
IN
2
2
1
1
=
1
Nello Sevastopoulos, et al. "How to Design High Order Filters with Stopband Notches Using the
LTC1562 Quad Operational Filter." Attached to this data sheet, available on the LTC web site
(www.linear-tech.com).
INV
V1
2nd ORDER
1/4 LTC1562
V2
R21
R
Q1
R
IN1
R
IN2
R
GAIN
I
O
R
FF2
C
IN1
V
IN
V
OUT
1562 F11
VIRTUAL
GROUND
+
Figure 11. Feedforward Notch Configuration for f
N
f
O
21
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
Note that the depth of the notch depends on the accuracy
of this resistor ratioing. The virtual-ground summing
point in Figure 11 may be from an op amp as shown, or in
a practical cascaded filter, the INV input of another Opera-
tional Filter block. The transfer function in Figure 11 with
C
IN1
= 0 is a "pure" notch (f
N
= f
0
) of the H
BR
(s) form above,
and the parameters are:
=
=
N
O
N
GAIN
FF
H
R
R
2
Because f
N
= f
0
in this case, the gain magnitude both at DC
and at high frequencies (f >> f
N
) is the same, H
N
(assuming
that the op amp in Figure 11 adds no significant frequency
response). Figure 12 shows this. Such a notch is ineffi-
cient as a cascaded part of a highpass, lowpass or bandpass
filter (the most common uses for notches). Variations of
Figure 11 can add a highpass or lowpass shape to the
notch, without using more Operational Filter blocks. The
key to doing so is to decouple the notch frequency f
N
from
the center frequency f
0
of the Operational Filter (this is
shown in Figures 13 and 15). The next two sections
summarize two variations of Figure 11 with this highpass/
lowpass shaping, and the remaining section shows a
different approach to building notches.
Feedforward Notches for f
N
> f
0
When C
IN1
0 in Figure 11, the notch frequency f
N
is above
the center frequency f
0
and the response has a lowpass
shape as well as a notch (Figure 13). C
IN1
contributes
phase lead, which increases the notch frequency above
the center frequency of the 2nd order Operational Filter
section. The resistor constraint from the previous section
also applies here and the H
BR
(s) parameters become:
=
=
N
O
IN
IN
Q
N
GAIN
FF
O
N
R C
R C
H
R
R
1
1
1
1
1
2
2
2
C is the internal capacitor value in the Operational Filter (in
the LTC1562, 159pF).
The configuration of Figure 11 is most useful for a stopband
notch in a lowpass filter or as an upper stopband notch in
a bandpass filter, since the two resistors R
IN2
and R
FF2
can
replace the input resistor R
IN
of either a lowpass section
(Figure 5) or a resistor-input bandpass section (Figure 6a)
built from a second Operational Filter. The configuration is
Figure 12. Notch Response with f
N
= f
O
FREQUENCY (kHz)
10
60
GAIN (dB)
40
20
20
100
1000
1562 F13
0
f
O
= 100kHz
f
N
= 200kHz
Q = 1
DC GAIN = 0dB
f
N
2
f
O
2
DC GAIN = H
N
( )
HIGH FREQ
GAIN = H
N
FREQUENCY (kHz)
10
100
GAIN (dB)
40
20
0
100
1000
AN54 TA18
60
80
f
N
= f
O
= 100kHz
H
N
= 1
Q = 1
Figure 13. Notch Response with f
N
> f
O
22
LTC1562
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
robust against tolerances in the C
IN1
value when f
N
ap-
proaches f
0
(for f
N
/f
0
1.4, as a rule of thumb) which is
attractive in narrow transition-band filters, because of the
relative cost of high accuracy capacitors. Further applica-
tion details appear in Part 1 of the series of articles.
1
Feedforward Notches for f
N
< f
0
Just as feedforward around an inverting bandpass section
yields a notch at the section's f
0
(Figure 11 with C
IN1
= 0),
feedforward around an inverting lowpass section causes
a notch at zero frequency (which is to say, a highpass
response). Moreover, and this is what makes it useful,
introducing a capacitor for phase lead moves the notch
frequency up from DC, exactly as C
IN1
in Figure 11 moves
the notch frequency up from the center frequency f
0
. In
Figure 14, the inverting lowpass output (V2) of the Opera-
tional Filter is summed, at a virtual ground, with a fed-
forward input signal. Capacitor C
IN1
shifts the resulting
notch frequency, f
N
, up from zero, giving a low frequency
notch with a highpass shape (Figure 15). The H
BR
(s)
response parameters are now:
=
=
N
O
Q
IN
IN
N
GAIN
FF
R
R
C
C
R
R
H
R
R
1
1
21
1
1
1
2
The constraint required for exact cancellation of the two
paths (i.e., for infinite notch depth) becomes:
R
R
R C
R C
IN
FF
Q
IN
2
2
1
1
1
=
R1 and C are the internal precision components (in the
LTC1562, 10k and 159pF respectively) as described above
in Setting f
0
and Q.
The configuration of Figure 14 is most useful as a lower
stopband notch in a bandpass filter, because the resistors
R
IN2
and R
FF2
can replace the input resistor R
IN
of a
bandpass section made from a second Operational Filter,
as in Figure 6a. The configuration is robust against toler-
ances in the C
IN1
value when f
N
approaches f
0
(for f
0
/f
N
1.4, as a rule of thumb) which is attractive in narrow
transition-band filters, because of the relative cost of high
accuracy capacitors. Further application details appear in
Part 2 of the series of articles.
1
FREQUENCY (Hz)
10k
60
GAIN (dB)
40
20
20
100k
1M
1562 F15
0
f
O
= 100kHz
f
N
= 50kHz
Q = 1
HIGH FREQ GAIN = 0dB
f
N
2
f
O
2
DC GAIN = H
N
( )
HIGH FREQ
GAIN = H
N
Figure 15. Notch Response with f
N
< f
0
Figure 14. Feedforward Notch Configuration for f
N
< f
O
INV
V1
2nd ORDER
1/4 LTC1562
V2
R21
R
Q1
R
IN1
R
IN2
R
GAIN
I
O
R
FF2
C
IN1
V
IN
V
OUT
1562 F14
VIRTUAL
GROUND
+
23
LTC1562
R-C Universal Notches
A different way to get 180
phase shift for a notch is to use
the built-in 90
phase difference between the two Opera-
tional Filter outputs along with a further 90
from an
external capacitor. This method achieves deep notches
independent of component matching, unlike the previous
techniques, and it is convenient for cascaded highpass as
well as lowpass and bandpass filters.
The V2 output of an Operational Filter is a time-integrated
version of V1 (see Figure 3), and therefore lags V1 by 90
over a wide range of frequencies. In Figure 16, a notch
response occurs when a 2nd order section drives a virtual-
ground input through two paths, one through a capacitor
and one through a resistor. Again, the virtual ground may
come from an op amp as shown, or from another Opera-
tional Filter's INV input. Capacitor C
N
adds a further 90
to
the 90
difference between V1 and V2, producing a
wideband 180
phase difference, but frequency-depen-
dent amplitude ratio, between currents I
R
and I
C
. At the
frequency where I
R
and I
C
have equal magnitude, I
O
becomes zero and a notch occurs. This gives a net transfer
function from V
IN
to V
OUT
in the form of H
BR
(s) as above,
with parameters:
=
=
N
N N
N
GAIN
IN
N
R C R C
H
R
R
C
C
1
2
1
1
APPLICATIO
N
S I
N
FOR
M
ATIO
N
W
U
U
U
DC Gain
R
R
R
R
High Frequency Gain
DC Gain
R C
R C
GAIN
IN
N
O
N
N N
=
=
=
1
2
2
21
21
R1 and C are the internal precision components (in the
LTC1562, 10k and 159pF respectively) as described above
in Setting f
0
and Q.
Unlike the notch methods of Figures 11 and 14, notch
depth from Figure 16 is inherent, not derived from compo-
nent matching. Errors in the R
N
or C
N
values alter the notch
frequency, f
N
, rather than the degree of cancellation at f
N
.
Also, the notch frequency, f
N
, is independent of the section's
center frequency f
0
, so f
N
can freely be equal to, higher
than or lower than f
0
(Figures 12, 13 or 15, respectively)
without changing the configuration. The chief drawback of
Figure 16 compared to the previous methods is a very
practical one--the C
N
capacitor value directly scales H
N
(and therefore the high frequency gain). Capacitor values
are generally not available in increments or tolerances as
fine as those of resistors, and this configuration lacks the
property of the previous two configurations that sensitiv-
ity to the capacitor value falls as f
N
approaches f
0
.
Figure 16. The R-C Universal Notch Configuration for an Operational Filter Block
INV
V1
2nd ORDER
1/4 LTC1562
V2
R21
R
Q1
R
IN1
R
N
R
GAIN
I
O
C
N
V
IN
V
OUT
1562 F16
VIRTUAL
GROUND
+
I
R
I
C
24
LTC1562
TYPICAL APPLICATIO
N
S
U
(Advanced)
R
Q1
30.1k
R
Q3
34k
R
IN3
31.6k
R
IN1
48.7k
V
IN
R
Q2
13k
R
IN2
37.4k
R
IN4
32.4k
C
IN2
24pF
R
Q4
11.5k
R22 57.6k
R24 32.4k
0.1
F
V
OUT
5V
5V
R21 31.6k
R23 31.6k
LTC1562
INVB
V1B
V2B
V
+
SHDN
V2A
V1A
INVA
20
19
18
16
15
13
12
11
INVC
V1C
V2C
V
AGND
V2D
V1D
INVD
1
2
3
5
6
8
9
10
0.1
F
C
IN4
10pF
1562 TA12a
C
IN3
18pF
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
8th Order 50kHz Lowpass Elliptic Filter
with 100dB Stopband Attenuation
8th Order 100kHz Elliptic Bandpass Filter
R
Q1
86.6k
R
Q3
71.5k
R
IN3
294k
C
IN3
18pF
R
IN1
95.3k
C
IN1
5.6pF
V
IN
R
Q2
84.5k
R
IN2
93.1k
R
FF2
301k
R
IN4
95.3k
R
FF4
332k
R
Q4
82.5k
R22 10k
R24 9.53k
0.1
F
V
OUT
5V
5V
R21 10.7k
R23 10k
LTC1562
INVB
V1B
V2B
V
+
SHDN
V2A
V1A
INVA
20
19
18
16
15
13
12
11
INVC
V1C
V2C
V
AGND
V2D
V1D
INVD
1
2
3
5
6
8
9
10
0.1
F
1562 F13a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
FREQUENCY (kHz)
USES THREE R-C UNIVERSAL NOTCHES AT f
N
= 133kHz, 167kHz, 222kHz.
DETAILED DESCRIPTION IN LINEAR TECHNOLOGY DESIGN NOTE 195.
WIDEBAND OUTPUT NOISE 60
V
RMS
120
60
80
100
20
0
20
40
1562 TA12b
GAIN (dB)
10
500
100
Amplitude Response
Amplitude Response
FREQUENCY (kHz)
25
90
GAIN (dB)
70
80
20
30
40
50
60
10
0
100
175
1562 TA13b
10
25
LTC1562
TYPICAL APPLICATIO
N
S
U
(Advanced)
R
Q1
95.3k
R
Q3
392k
R
IN1B
69.8k
R
IN1A
140k
V
IN
R
Q2
182k
R
IN3
536k
C
IN3
27pF
C
IN2
33pF
R
IN2
249k
R
IN4
301k
C
IN4
56pF
R
Q4
66.5k
R22 226k
R24 649k
0.1
F
V
OUT
V
+
V
V
TO
PIN 10
R21 324k
R23 196k
LTC1562
INVB
V1B
V2B
V
+
SHDN
V2A
V1A
INVA
20
19
18
16
15
13
12
11
INVC
V1C
V2C
V
AGND
V2D
V1D
INVD
1
2
3
5
6
8
9
10
0.1
F
C
IN1
390pF
1562 F14a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
9th Order 22kHz Lowpass Elliptic Filter
FREQUENCY (kHz)
5
90
GAIN (dB)
70
50
30
10
0
80
60
40
20
10
50
1562 TA14b
10
Amplitude Response
FREQUENCY (kHz)
1
90
NOISE + THD (dB)
80
70
60
50
10
20
1562 TA14c
40
85
75
65
55
45
V
IN
= 1.65V
RMS
= 4.6V
P-P
V
S
=
5V
Noise + THD vs Frequency
26
LTC1562
Dual 5th Order Lowpass "Elliptic" Filter
R
Q1
R
Q1
R
IN1B
R
IN1A
V
IN1
V
IN2
R
Q2
C
IN2
R
IN2
R
IN2
C
IN2
R
Q2
R22
R22
0.1
F
V
OUT2
V
OUT1
5V
5V
R21
R21
LTC1562
INVB
V1B
V2B
V
+
SHDN
V2A
V1A
INVA
20
19
18
16
15
13
12
11
INVC
V1C
V2C
V
AGND
V2D
V1D
INVD
1
2
3
5
6
8
9
10
0.1
F
C
IN1
R
IN1B
R
IN1A
C
IN1
1562 TA15a
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
Construction and Instrumentation Cautions
100dB rejections at hundreds of kilohertz require electri-
cally clean, compact construction, with good grounding
and supply decoupling, and minimal parasitic capaci-
tances in critical paths (such as Operational Filter INV
inputs). In a circuit with 5k resistances trying for 100dB
rejection at 100kHz, a stray coupling of 0.003pF around
the signal path can preclude the 100dB. (By comparison,
the stray capacitance between two adjacent pins of an IC
can be 1pF or more.) Also, high quality supply bypass
capacitors of 0.1
F near the chip provide good decoupling
from a clean, low inductance power source. But several
inches of wire (i.e., a few microhenrys of inductance) from
the power supplies, unless decoupled by substantial
capacitance (
10
F) near the chip, can cause a high-Q LC
resonance in the hundreds of kHz in the chip's supplies or
ground reference, impairing stopband rejection and other
specifications at those frequencies. In demanding filter
circuits we have often found that a compact, carefully laid
out printed circuit board with good ground plane makes a
difference of 20dB in both stopband rejection and distor-
tion performance. Highly selective circuits can even ex-
hibit these issues at frequencies well below 100kHz.
Finally, equipment to measure filter performance can itself
introduce distortion or noise floors; checking for these
limits with a wire replacing the filter is a prudent routine
procedure.
f
C
(Hz)
R
IN1A
R
IN1B
C
IN1
R
Q1
R21
R
IN2
C
IN2
R
Q2
R22
100k
5.9k
7.5k
680pF
28k
7.5k
6.34k
68pF
9.31k
11.3k
75k
8.06k
15.4k
560pF
36.5k
13.3k
11.3k
68pF
12.7k
20k
50k
16.9k
35.7k
390pF
56.2k
30.1k
25.5k
68pF
18.7k
44.2k
Amplitude Response
FREQUENCY (kHz)
120
60
80
100
20
0
20
40
1562 TA15b
GAIN (dB)
10
1000
100
f
C
= 100kHz
TYPICAL APPLICATIO
N
S
U
(Advanced)
27
LTC1562
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no represen-
tation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
Dimensions in inches (millimeters) unless otherwise noted.
PACKAGE DESCRIPTIO
N
U
G Package
20-Lead Plastic SSOP (0.209)
(LTC DWG # 05-08-1640)
G20 SSOP 0595
0.005 0.009
(0.13 0.22)
0
8
0.022 0.037
(0.55 0.95)
0.205 0.212**
(5.20 5.38)
0.301 0.311
(7.65 7.90)
1
2 3
4
5
6 7 8
9 10
0.278 0.289*
(7.07 7.33)
17
18
14 13 12 11
15
16
19
20
0.068 0.078
(1.73 1.99)
0.002 0.008
(0.05 0.21)
0.0256
(0.65)
BSC
0.010 0.015
(0.25 0.38)
DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH
SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD
FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
*
**
28
LTC1562
1562f LT/TP 0199 4K PRINTED IN USA
LINEAR TECHNOLOGY CORPORATION 1998
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900
q
FAX: (408) 434-0507
q
www.linear-tech.com
TYPICAL APPLICATIO
N
U
Amplitude Response
FREQUENCY (kHz)
1
80
GAIN (dB)
20
0
20
10
100
300
1562 TA16b
40
60
f
C
= 64kHz
f
C
= 16kHz
f
C
= 32kHz
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1068, LTC1068-X
Quad 2-Pole Switched Capacitor Building Block Family
Clock-Tuned
LTC1560-1
5-Pole Elliptic Lowpass, f
C
= 1MHz/0.5MHz
No External Components, SO8
LTC1562-2
Quad 2-Pole Active RC, 20kHz to 300kHz
Same Pinout as the LTC1562
f
C
(Hz)
R
IN1
= R
IN3
R21 = R23
R
Q1
= R
Q3
R
IN2
= R
IN4
R22 = R24
R
Q2
= R
Q4
16k
105k
105k
34k
340k
340k
34k
32k
26.1k
26.1k
16.9k
84.5k
84.5k
16.9k
64k
8.45k
6.49k
8.45k
16.2k
21k
8.45k
1V/DIV
10
s/DIV
1562 TA16c
Dual 4th Order 12dB Gaussian Lowpass Filter
1
2
3
5
6
8
9
10
20
19
18
16
15
13
12
11
INV B
V1 B
V2 B
V
+
SHDN
V2 A
V1 A
INV A
INV C
V1 C
V2 C
V
AGND
V2 D
V1 D
INV D
LTC1562
R
IN2
R
IN4
R
IN1
V
IN2
V
IN1
SCHEMATIC INCLUDES PIN NUMBERS FOR 20-PIN PACKAGE.
PINS 4, 7, 14, 17 (NOT SHOWN) ALSO CONNECT TO V
V
OUT1
1562 TA16a
V
OUT2
R
IN3
5V
R
Q1
R21
R23
0.1
F
1
F
R
Q3
R24
R
Q4
R
Q2
R22
4-Level Eye Diagram
f
C
= 16kHz, Data Clock = 32kHz
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