LM4733
3 Channel 30W Audio Power Amplifier with Mute

General Description

The LM4733 is a three channel audio amplifier capable of
typically delivering 30W per channel of continuous average
output power into a 4

or 8 load with less than 10%

THD+N from 20Hz - 20kHz.

The LM4733 has short circuit protection and a thermal shut
down feature that is activated when the die temperature
exceeds 150°C. The LM4733 also has an under voltage lock
out feature for click and pop free power on and off.

Each amplifier of the LM4733 has an independent smooth
transition fade-in/out mute.

The LM4733 has a wide operating supply range from

10V

32V allowing for lower cost unregulated power supplies to

be used.

The LM4733 amplifiers can easily be configured for bridge or
parallel operation for higher power and bi-amp solutions

Key Specifications

Output Power/Channel at 10% THD+N,

1kHz into 4

or 8

30W (typ)

THD+N at 3 x 1W into 8

, 1kHz

0.03% (typ)

Mute Attenuation

110dB (typ)

PSRR

85dB (typ)

Slew Rate

9V/µs (typ)

Features

Low external component count

Quiet fade-in/out mute mode

Wide supply range: 20V - 64V

Applications

Audio amplifier for component stereo

Audio amplifier for compact stereo

Audio amplifier for self-powered speakers

Audio amplifier for high-end and HD TVs

Typical Application

200794B5

FIGURE 1. Typical Audio Amplifier Application Circuit

September 2003

LM4733

Channel

30W

Audio

Power

Amplifier

with

Mute

DS200794

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Absolute Maximum Ratings

(Notes 1,

If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.

Supply Voltage |V

| + |V

69V

Common Mode Input Voltage

or V

) and

| + |V

60V

Differential Input Voltage (Note 12)

60V

Output Current

Internally Limited

Power Dissipation (Note 3)

125W

ESD Susceptability (Note 4)

2.0kV

ESD Susceptability (Note 5)

200V

Junction Temperature (T

JMAX

) (Note 9)

150°C

Soldering Information

TA Package (10 seconds)

260°C

Storage Temperature

-40°C to +150°C

Thermal Resistance

30°C/W

0.9°C/W

Operating Ratings

(Notes 1, 2)

Temperature Range

MIN

MAX

-20°C

+85°C

Supply Voltage |V

| + |V

20V

TOTAL

64V

Electrical Characteristics

(Notes 1, 2)

The following specifications apply for V

= +24V, V

= -24V, I

MUTE

= -1mA/channel and R

= 8

unless otherwise specified.

Limits apply for T

= 25°C.

Symbol

Parameter

Conditions

LM4733

Units

(Limits)

Typical

Limit

(Note 6)

(Notes 7, 8)

| + |V

Power Supply Voltage (Note

10)

GND - V

V (min)

V (max)

Mute Attenuation

MUTE

= 0mA

110

Output Power (RMS)

THD+N = 10% (max), f = 1kHz,

| = |V

| = 19V, R

= 4

| = |V

| = 24V, R

= 8

W (min)

THD+N = 1% (max), f = 1kHz,

| = |V

| = 19V, R

= 4

| = |V

| = 24V, R

= 8

THD+N

Total Harmonic Distortion +

Noise

= 1W, f = 1kHz

= 26dB

| = |V

| = 19V, R

= 4

| = |V

| = 24V, R

= 8

0.05

0.03

talk

Channel Separation (Note 11)

= 10W, f = 1kHz

= 10W, f = 10kHz

Slew Rate

= 1.2V

RMS

, f = 10kHz square

Wave, R

= 2k

V/µs

Total Quiescent Power

= 0V,

150

mA (max)

Supply Current

= 0V, I

= 0A

Input Offset Voltage

= 0V, I

= 0mA

mV (max)

Input Bias Current

= 0V, I

= 0mA

0.2

µA

PSRR

Power Supply Rejection Ratio

= 24V + V

RIPPLE

(1V

RMS

)

RIPPLE

= 120Hz sine, V

= -24V

= -24V + V

RIPPLE

(1V

RMS

)

RIPPLE

= 120Hz sine, V

= 24V

VOL

Open Loop Voltage Gain

| = |V

| = 24V, R

= 2k

= 20V

115

Input Noise

IHF-A-Weighting Filter,

= 600

(Input Referred)

3.0

µV

Note 1: All voltages are measured with respect to the ground pins, unless otherwise specified.

LM4733

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Electrical Characteristics

(Notes 1, 2) (Continued)

Note 2: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
functional, but do not guarantee specific performance limits. Electrical Characteristics state DC and AC electrical specifications under particular test conditions which
guarantee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit
is given; however, the typical value is a good indication of device performance.

Note 3: The maximum power dissipation must be de-rated at elevated temperatures and is dictated by T

JMAX

, and the ambient temperature T

. The maximum

allowable power dissipation is P

DMAX

= (T

JMAX

-T

or the number given in the Absolute Maximum Ratings, whichever is lower. For the LM4733, T

JMAX

= 150°C

and the typical

is 0.9°C/W. Refer to the DETERMINING THE CORRECT HEAT SINK section for more information.

Note 4: Human body model, 100pF discharged through a 1.5k

resistor.

Note 5: Machine Model: a 220pF - 240pF discharged through all pins.

Note 6: Typical specifications are measured at 25°C and represent the parametric norm.

Note 7: Tested limits are guaranteed to National's AOQL (Average Outgoing Quality Level).

Note 8: Datasheet min/max specification limits are guaranteed by design, test, or statistical analysis.

Note 9: The maximum operating junction temperature is 150°C. However, the instantaneous Safe Operating Area temperature is 250°C.

Note 10: V

must have at least - 9V at its pin with reference to GND in order for the under-voltage protection circuitry to be disabled. In addition, the voltage

differential between V

and V

must be greater than 14V.

Note 11: Cross talk performance was measured using the demo board shown in the datasheet. PCB layout will affect cross talk. It is recommended that the input
and output traces be separated by as much distance as possible. Return ground traces from outputs should also be independent back to sinlge ground point and
use as wide of traces as possible.

Note 12: The Differential Input Voltage Absolute Maximum Rating is based on supply voltages V

= 30V and V

= - 30V.

Note 13: The TA27A is a non-isolated package. The package's metal back and any heat sink to which it is mounted are connected to the V

potential when using

only thermal compound. If a mica washer is used in addition to thermal compound,

(case to sink) is increased, but the heat sink will be electrically isolated from

LM4733

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External Components Description

(Figures 1-5)

Components

Functional Description

Prevents current from entering the amplifier's non-inverting input. This current may pass through to the load

during system power down, because of the amplifier's low input impedance when the undervoltage circuitry

is off. This phenomenon occurs when the V

and V

supply voltages are below 1.5V.

Inverting input resistance. Along with R

, sets AC gain.

Feedback resistance. Along with R

, sets AC gain.

(Note 14)

Feedback resistance. Works with Cf and Rf creating a lowpass filter that lowers AC gain at high

frequencies. The -3dB point of the pole occurs when: (R

- R

)/2 = R

// [1/(2

) + R

] for the

Non-Inverting configuration shown in Figure 5.

(Note 14)

Compensation capacitor. Works with R

and R

to reduce AC gain at higher frequencies.

(Note 14)

Compensation capacitor. Reduces the gain at higher frequencies to avoid quasi-saturation oscillations of the

output transistor. Also suppresses external electromagnetic switching noise created from fluorescent lamps.

(Note 14)

Feedback capacitor which ensures unity gain at DC. Along with R

also creates a highpass filter at f

1/(2

Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for proper

placement and selection of bypass capacitors.

(Note 14)

Acts as a volume control by setting the input voltage level.

(Note 14)

Sets the amplifier's input terminals DC bias point when C

is present in the circuit. Also works with C

create a highpass filter at f

= 1/(2

). If the value of R

is too large, oscillations may be observed on

the outputs when the inputs are floating. Recommended values are 10k

to 47k. Refer to Figure 5.

(Note 14)

Input capacitor. Prevents the input signal's DC offsets from being passed onto the amplifier's inputs.

(Note 14)

Works with C

to stabilize the output stage by creating a pole that reduces high frequency instabilities.

(Note 14)

Works with R

to stabilize the output stage by creating a pole that reduces high frequency instabilities. The

pole is set at f

= 1/(2

). Refer to Figure 5.

L (Note 14)

Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce

the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and

pass audio signals to the load. Refer to Figure 5.

R (Note 14)

Provides DC voltage biasing for the transistor Q1 in single supply operation.

Provides bias filtering for single supply operation.

INP

(Note 14)

Limits the voltage difference between the amplifier's inputs for single supply operation. Refer to the Clicks

and Pops application section for a more detailed explanation of the function of R

INP

Provides input bias current for single supply operation. Refer to the Clicks and Pops application section for

a more detailed explanation of the function of R

Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the

half-supply point along with C

Mute resistance set up to allow 0.5mA to be drawn from each MUTE pin to turn the muting function off.
R

is calculated using: R

(|V

| - 2.6V)/l where l

0.5mA. Refer to the Mute Attenuation vs Mute

Current curves in the Typical Performance Characteristics section.

Mute capacitance set up to create a large time constant for turn-on and turn-off muting.

Mute switch. When open or switched to GND, the amplifier will be in mute mode.

OUT

Reduces current flow between outputs that are caused by Gain or DC offset differences between the

amplifiers.

Note 14: Optional components dependent upon specific design requirements.

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Typical Performance Characteristics

(Continued)

Output Power vs Supply Voltage

Chs 1&3 in Bridge Mode (Note15),

f = 1kHz, R

= 8

, 80kHz BW

Output Power vs Supply Voltage

Chs 1&3 in Parallel Mode (Note16),

f = 1kHz, R

= 4

, 80kHz BW

20079413

20079415

Output Power vs Supply Voltage

All Chs in Parallel Mode (Note16),

f = 1kHz, R

= 2

, 80kHz BW

20079414

Note 15: Bridge mode graphs were taken using the demo board and inverting the signal to the channel B input.

Note 16: Parallel mode graphs were taken using the demo board connecting each output through a 0.1

/3W resistor to the load.

Application Information

MUTE MODE

The muting function allows the user to mute the amplifier.
This can be accomplished as shown in the Typical Applica-
tion Circuit. The resistor R

is chosen with reference to the

negative supply voltage and is used in conjunction with a
switch. The switch, when opened or switched to GND, cuts
off the current flow from the MUTE pins to -V

, thus placing

the LM4733 into mute mode. Refer to the Mute Attenuation
vs Mute Current curves in the Typical Performance Char-
acteristics section for values of attenuation per current out
of each MUTE pin. The resistance R

is calculated by the

following equation:

(|-V

| - 2.6V) / I

MUTE

Where I

MUTE

0.5mA for each MUTE pin.

The MUTE pins can be tied together so that only one resistor
is required for the mute function. The mute resistor value

must be chosen so that a minimum of 1.5mA is pulled
through the resistor R

. This ensures that each amplifier is

fully operational. Taking into account supply line fluctuations,
it is a good idea to pull out 1mA per MUTE pin or 3mA total
if all pins are tied together.

A turn-on MUTE or soft start circuit may also be used during
power up. A simple circuit like the one shown below may be
used.

200794B7

The RC combination of C

and R

may cause the voltage

at point A to change more slowly than the -V

supply

LM4733

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Application Information

(Continued)

voltage. Until the voltage at point A is low enough to have
0.5mA of current per MUTE pin flow through R

, the IC will

be in mute mode. The series combination of R

and R

needs to satisfy the mute equation above for all operating
voltages or mute mode may be activated during normal
operation. For a longer turn-on mute time, a larger time
constant,

= RC = R

(sec), is needed. For the values

show above and with the MUTE pins tied together, the
LM4733 will enter play mode when the voltage at point A is
-25.1V. The voltage at point A is found with Equation (1)
below.

(t) = (V

- V

-t/

(Volts)

(1)

where:

t = time (sec)
= RC (sec)
V

= Voltage on C at t = 0 (Volts)

= Final voltage, -V

in this circuit (Volts)

UNDER-VOLTAGE PROTECTION

Upon system power-up, the under-voltage protection cir-
cuitry allows the power supplies and their corresponding
capacitors to come up close to their full values before turning
on the LM4733. Since the supplies have essentially settled
to their final value, no DC output spikes occur. At power
down, the outputs of the LM4733 are forced to ground before
the power supply voltages fully decay preventing transients
on the output.

OVER-VOLTAGE PROTECTION

The LM4733 contains over-voltage protection circuitry that
limits the output current while also providing voltage clamp-
ing. The clamp does not, however, use internal clamping
diodes. The clamping effect is quite the same because the
output transistors are designed to work alternately by sinking
large current spikes.

THERMAL PROTECTION

The LM4733 has a sophisticated thermal protection scheme
to prevent long-term thermal stress of the device. When the
temperature on the die exceeds 150°C, the LM4733 shuts
down. It starts operating again when the die temperature
drops to about 145°C, but if the temperature again begins to
rise, shutdown will occur again above 150°C. Therefore, the
device is allowed to heat up to a relatively high temperature
if the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion be-
tween the thermal shutdown temperature limits of 150°C and
145°C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.

Since the die temperature is directly dependent upon the
heat sink used, the heat sink should be chosen so that
thermal shutdown is not activated during normal operation.
Using the best heat sink possible within the cost and space
constraints of the system will improve the long-term reliability
of any power semiconductor device, as discussed in the
Determining the Correct Heat Sink section.

DETERMlNlNG MAXIMUM POWER DISSIPATION

Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understand-
ing if optimum power output is to be obtained. An incorrect

maximum power dissipation calculation may result in inad-
equate heat sinking causing thermal shutdown and thus
limiting the output power.

Equation (2) shows the theoretical maximum power dissipa-
tion point for each amplifier in a single-ended configuration
where V

is the total supply voltage.

DMAX

= (V

)

/ 2

(2)

Thus by knowing the total supply voltage and rated output
load, the maximum power dissipation point can be calcu-
lated. The package dissipation is three times the number
which results from Equation (2) since there are three ampli-
fiers in each LM4733. Refer to the graphs of Power Dissipa-
tion versus Output Power in the Typical Performance Char-
acteristics section which show the actual full range of power
dissipation not just the maximum theoretical point that re-
sults from Equation (2).

DETERMINING THE CORRECT HEAT SINK

The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such
that the thermal protection circuitry is not activated under
normal circumstances.

The thermal resistance from the die to the outside air,

(junction to ambient), is a combination of three thermal re-
sistances,

(junction to case),

(case to sink), and

(sink to ambient). The thermal resistance,

(junction to

case), of the LM4733T is 0.9°C/W. Using Thermalloy Ther-
macote thermal compound, the thermal resistance,

(case to sink), is about 0.2°C/W. Since convection heat flow
(power dissipation) is analogous to current flow, thermal
resistance is analogous to electrical resistance, and tem-
perature drops are analogous to voltage drops, the power
dissipation out of the LM4733 is equal to the following:

DMAX

= (T

JMAX

-T

AMB

) /

(3)

where T

JMAX

= 150°C, T

AMB

is the system ambient tempera-

ture and

200794B8

Once the maximum package power dissipation has been
calculated using Equation 2, the maximum thermal resis-
tance,

, (heat sink to ambient) in °C/W for a heat sink can

be calculated. This calculation is made using Equation 4
which is derived by solving for

in Equation 3.

= [(T

JMAX

-T

AMB

)-P

DMAX

(

)] / P

DMAX

(4)

Again it must be noted that the value of

is dependent

upon the system designer's amplifier requirements. If the
ambient temperature that the audio amplifier is to be working
under is higher than 25°C, then the thermal resistance for the
heat sink, given all other things are equal, will need to be
smaller.

LM4733

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Application Information

(Continued)

SUPPLY BYPASSING

The LM4733 has excellent power supply rejection and does
not require a regulated supply. However, to improve system
performance as well as eliminate possible oscillations, the
LM4733 should have its supply leads bypassed with low-
inductance capacitors having short leads that are located
close to the package terminals. Inadequate power supply
bypassing will manifest itself by a low frequency oscillation
known as "motorboating" or by high frequency instabilities.
These instabilities can be eliminated through multiple by-
passing utilizing a large tantalum or electrolytic capacitor
(10µF or larger) which is used to absorb low frequency
variations and a small ceramic capacitor (0.1µF) to prevent
any high frequency feedback through the power supply lines.

If adequate bypassing is not provided, the current in the
supply leads which is a rectified component of the load
current may be fed back into internal circuitry. This signal
causes distortion at high frequencies requiring that the sup-
plies be bypassed at the package terminals with an electro-
lytic capacitor of 470µF or more.

BRIDGED AMPLIFIER APPLICATION

The LM4733 has three operational amplifiers internally, al-
lowing for a few different amplifier configurations. One of
these configurations is referred to as "bridged mode" and
involves driving the load differentially through two of the
LM4733's outputs. This configuration is shown in Figure 2.
Bridged mode operation is different from the classical single-
ended amplifier configuration where one side of its load is
connected to ground.

A bridge amplifier design has a distinct advantage over the
single-ended configuration, as it provides differential drive to
the load, thus doubling output swing for a specified supply
voltage. Theoretically, four times the output power is pos-
sible as compared to a single-ended amplifier under the
same conditions. This increase in attainable output power
assumes that the amplifier is not current limited or clipped.

A direct consequence of the increased power delivered to
the load by a bridge amplifier is an increase in internal power
dissipation. For each operational amplifier in a bridge con-
figuration, the internal power dissipation will increase by a
factor of two over the single ended dissipation. Using Equa-
tion (2) the load impedance should be divided by a factor of
two to find the maximum power dissipation point for each
amplifier in a bridge configuration. In the case of an 8

load

in a bridge configuration, the value used for R

in Equation

(2) would be 4

for each amplifier in the bridge. When using

two of the amplifiers of the LM4733 in bridge mode, the third
amplifier should have a load impedance equal to or higher
than the equivalent impedance seen by each of the bridged
amplifiers. In the example above where the bridge load is 8

and each amplifier in the bridge sees a load value of 4

then

the third amplifier should also have a 4

load impedance or

higher. Using a lower load impedance on the third amplifier
will result in higher power dissipation in the third amplifier
than the other two amplifiers and may result in unwanted
activation of thermal shut down on the third amplifier. Once
the impedance seen by each amplifier is known then Equa-
tion (2) can be used to calculated the value of P

DMAX

for

each amplifier. The P

DMAX

of the IC package is found by

adding up the power dissipation for each amplifier within the
IC package.

This value of P

DMAX

can be used to calculate the correct size

heat sink for a bridged amplifier application. Since the inter-

nal dissipation for a given power supply and load is in-
creased by using bridged-mode, the heatsink's

will have

to decrease accordingly as shown by Equation 4. Refer to
the section, Determining the Correct Heat Sink, for a more
detailed discussion of proper heat sinking for a given appli-
cation.

PARALLEL AMPLIFIER APPLICATION

Parallel configuration is normally used when higher output
current is needed for driving lower impedance loads (i.e. 4

or lower) to obtain higher output power levels. As shown in
Figure 3 , the parallel amplifier configuration consist of de-
signing the amplifiers in the IC to have identical gain, con-
necting the inputs in parallel and then connecting the outputs
in parallel through a small external output resistor. Any num-
ber of amplifiers can be connected in parallel to obtain the
needed output current or to divide the power dissipation
across multiple IC packages. Ideally, each amplifier shares
the output current equally. Due to slight differences in gain
the current sharing will not be equal among all channels. If
current is not shared equally among all channels then the
power dissipation will also not be equal among all channels.
It is recommended that 0.1% tolerance resistors be used to
set the gain (R

and R

) for a minimal amount of difference in

current sharing.

When operating two or more amplifiers in parallel mode the
impedance seen by each amplifier is equal to the total load
impedance multiplied by the number of amplifiers driving the
load in parallel as shown by Equation (5) below:

L(parallel)

= R

L(total)

x Number of amplifiers

(5)

Once the impedance seen by each amplifier in the parallel
configuration is known then Equation (2) can be used with
this calculated impedance to find the amount of power dis-
sipation for each amplifier. Total power dissipation (P

DMAX

)

within an IC package is found by adding up the power
dissipation for each amplifier in the IC package. Using the
calculated P

DMAX

the correct heat sink size can be deter-

mined. Refer to the section, Determining the Correct Heat
Sink, for more information and detailed discussion of proper
heat sinking.

If only two amplifiers of the LM4733 are used in parallel
mode then the third amplifier should have a load impedance
equal to or higher than the equivalent impedance seen by
each of the amplifiers in parallel mode. Having the same
load impedance on all amplifiers means that the power
dissipation in each amplifier will be equal. Using a lower load
impedance on the third amplifier will result in higher power
dissipation in the third amplifier than the other two amplifiers
and may result in unwanted activation of thermal shut down
on the third amplifier. Having a higher impedance on the third
amplifier than the equivalent impedance on the two amplifi-
ers in parallel will reduce total IC package power dissipation
reducing the heat sink size requirement.

BI-AMP AND TRI-AMP APPLICATIONS

Bi-amping is the practice of using two different amplifiers to
power the individual drivers in a speaker enclosure. For
example, a two-way speaker enclosure might have a tweeter
and a subwoofer. One amplifier would drive the tweeter and
another would drive the subwoofer. One advantage is that
the gain of each amplifier can be adjusted for the different
driver sensitivities. Another advantage is the crossover can
be designed before the amplifier stages with low cost op
amps instead of large passive components. With the cross-
over before the amplifier stages no power is wasted in the
passive crossover as each individual amplifier provides the

LM4733

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Application Information

(Continued)

correct frequencies for the driver. Tri-Amping is using three
different amplifier stages in the same way bi-amping is done.
Bi-amping can also be done on a three-way speaker design
by using one amplifier for the subwoofer and another for the
midrange and tweeter.

The LM4733 is perfectly suited for bi-amp or tri-amp appli-
cations with it's three amplifiers. Two of the amplifiers can be
configured for bridge or parallel mode to drive a subwoofer
with the third amplifier driving the tweeter or tweeter and
midrange. An example would be to use a 4

subwoofer and

tweeter/midrange with the LM4733 in parallel and single-

ended modes. Each amplifier would see an 8

load but the

subwoofer would have twice the output power as the
tweeter/midrange. The gain of each amplifier may also be
adjusted for the desired response. Using the LM4733 in a
tri-amp configuration would allow the gain of each amplifier
to be adjusted to achieve the desired speaker response.

SINGLE-SUPPLY AMPLIFIER APPLICATION

The typical application of the LM4733 is a split supply am-
plifier. But as shown in Figure 4, the LM4733 can also be
used in a single power supply configuration. This involves
using some external components to create a half-supply bias
which is used as the reference for the inputs and outputs.
Thus, the signal will swing around half-supply much like it
swings around ground in a split-supply application. Along
with proper circuit biasing, a few other considerations must
be accounted for to take advantage of all of the LM4733
functions, like the mute function.

CLICKS AND POPS

In the typical application of the LM4733 as a split-supply
audio power amplifier, the IC exhibits excellent "click" and
"pop" performance when utilizing the mute mode. In addition,
the device employs Under-Voltage Protection, which elimi-
nates unwanted power-up and power-down transients. The
basis for these functions are a stable and constant half-
supply potential. In a split-supply application, ground is the
stable half-supply potential. But in a single-supply applica-
tion, the half-supply needs to charge up at the same rate as
the supply rail, V

. This makes the task of attaining a

clickless and popless turn-on more challenging. Any uneven
charging of the amplifier inputs will result in output clicks and
pops due to the differential input topology of the LM4733.

To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM4733. In Figure 4, the resistor R

INP

serves to keep the inputs at the same potential by limiting the
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
based on a specific application loading and thus, the system
designer may need to adjust these values for optimal perfor-
mance.

As shown in Figure 4, the resistors labeled R

help bias up

the LM4733 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of R

, namely 10k

and 200k. These

resistors bring up the inputs at the same rate resulting in a
popless turn-on. Adjusting these resistors values slightly
may reduce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.

PROPER SELECTION OF EXTERNAL COMPONENTS

Proper selection of external components is required to meet
the design targets of an application. The choice of external
component values that will affect gain and low frequency
response are discussed below.

The gain of each amplifier is set by resistors R

and R

for the

non-inverting configuration shown in Figure 1. The gain is
found by Equation (6) below:

= 1 + R

/ R

(V/V)

(6)

For best noise performance, lower values of resistors are
used. A value of 1k

is commonly used for R

and then

setting the value of R

for the desired gain. For the LM4733

the gain should be set no lower than 10V/V and no higher
than 50V/V. Gain settings below 10V/V may experience
instability and using the LM4733 for gains higher than 50V/V
will see an increase in noise and THD.

The combination of R

with C

(see Figure 1) creates a high

pass filter. The low frequency response is determined by
these two components. The -3dB point can be found from
Equation (7) shown below:

= 1 / (2

) (Hz)

(7)

If an input coupling capacitor is used to block DC from the
inputs as shown in Figure 5, there will be another high pass
filter created with the combination of C

and R

. When

using a input coupling capacitor R

is needed to set the DC

bias point on the amplifier's input terminal. The resulting
-3dB frequency response due to the combination of C

and

can be found from Equation (8) shown below:

= 1 / (2

) (Hz)

(8)

With large values of R

oscillations may be observed on the

outputs when the inputs are left floating. Decreasing the
value of R

or not letting the inputs float will remove the

oscillations. If the value of R

is decreased then the value of

will need to increase in order to maintain the same -3dB

frequency response.

HIGH PERFORMANCE CONSIDERATIONS

Using low cost electrolytic capacitors in the signal path such
as C

and C

(see Figures 1 - 5) will result in very good

performance. However, electrolytic capacitors are less linear
than other premium capacitors. Higher THD+N performance
may be obtained by using high quality polypropylene capaci-
tors in the signal path. A more cost effective solution may be
the use of smaller value premium capacitors in parallel with
the larger electrolytic capacitors. This will maintain signal
quality in the upper audio band where any degradation is
most noticeable while also coupling in the signals in the
lower audio band for good bass response.

Distortion is introduced as the audio signal approaches the
lower -3dB point, determined as discussed in the section
above. By using larger values of capacitors such that the
-3dB point is well outside of the audio band will reduce this
distortion and improve THD+N performance.

Increasing the value of the large supply bypass capacitors
will improve burst power output. The larger the supply by-
pass capacitors the higher the output pulse current without
supply droop increasing the peak output power. This will also
increase the headroom of the amplifier and reduce THD.

SIGNAL-TO-NOISE RATIO

In the measurement of the signal-to-noise ratio, misinterpre-
tations of the numbers actually measured are common. One
amplifier may sound much quieter than another, but due to
improper testing techniques, they appear equal in measure-

LM4733

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Application Information

(Continued)

ments. This is often the case when comparing integrated
circuit designs to discrete amplifier designs. Discrete transis-
tor amps often "run out of gain" at high frequencies and
therefore have small bandwidths to noise as indicated below.

20079499

Integrated circuits have additional open loop gain allowing
additional feedback loop gain in order to lower harmonic
distortion and improve frequency response. It is this addi-
tional bandwidth that can lead to erroneous signal-to-noise
measurements if not considered during the measurement
process. In the typical example above, the difference in
bandwidth appears small on a log scale but the factor of 10in
bandwidth, (200kHz to 2MHz) can result in a 10dB theoreti-
cal difference in the signal-to-noise ratio (white noise is
proportional to the square root of the bandwidth in a system).

In comparing audio amplifiers it is necessary to measure the
magnitude of noise in the audible bandwidth by using a
"weighting" filter (Note 17). A "weighting" filter alters the
frequency response in order to compensate for the average
human ear's sensitivity to the frequency spectra. The weight-
ing filters at the same time provide the bandwidth limiting as
discussed in the previous paragraph.

Note 17: CCIR/ARM: A Practical Noise Measurement Method; by Ray
Dolby, David Robinson and Kenneth Gundry, AES Preprint No. 1353 (F-3).

In addition to noise filtering, differing meter types give differ-
ent noise readings. Meter responses include:

RMS reading,

average responding,

peak reading, and

quasi peak reading.

Although theoretical noise analysis is derived using true
RMS based calculations, most actual measurements are
taken with ARM (Average Responding Meter) test equip-
ment.

Typical signal-to-noise figures are listed for an A-weighted
filter which is commonly used in the measurement of noise.
The shape of all weighting filters is similar, with the peak of
the curve usually occurring in the 3kHz7kHz region.

LEAD INDUCTANCE

Power op amps are sensitive to inductance in the output
leads, particularly with heavy capacitive loading. Feedback
to the input should be taken directly from the output terminal,
minimizing common inductance with the load.

Lead inductance can also cause voltage surges on the sup-
plies. With long leads to the power supply, energy is stored in
the lead inductance when the output is shorted. This energy
can be dumped back into the supply bypass capacitors when
the short is removed. The magnitude of this transient is
reduced by increasing the size of the bypass capacitor near
the IC. With at least a 20µF local bypass, these voltage
surges are important only if the lead length exceeds a couple
feet (

1µH lead inductance). Twisting together the supply

and ground leads minimizes the effect.

PHYSICAL IC MOUNTING CONSIDERATIONS

Mounting of the package to a heat sink must be done such
that there is sufficient pressure from the mounting screws to
insure good contact with the heat sink for efficient heat flow.
Over tightening the mounting screws will cause the package
to warp reducing contact area with the heat sink. Less
contact with the heat sink will increase the thermal resis-
tance from the package case to the heat sink (

) resulting

in higher operating die temperatures and possible unwanted
thermal shut down activation. Extreme over tightening of the
mounting screws will cause severe physical stress resulting
in cracked die and catastrophic IC failure. The recom-
mended mounting screw size is M3 with a maximum torque
of 50 N-cm. Additionally, it is best to use washers under the
screws to distribute the force over a wider area or a screw
with a wide flat head. To further distribute the mounting force
a solid mounting bar in front of the package and secured in
place with the two mounting screws may be used. Other
mounting options include a spring clip. If the package is
secured with pressure on the front of the package the maxi-
mum pressure on the molded plastic should not exceed
150N/mm

Additionally, if the mounting screws are used to force the
package into correct alignment with the heat sink, package
stress will be increased. This increase in package stress will
result in reduced contact area with the heat sink increasing
die operating temperature and possible catastrophic IC fail-
ure.

LM4733

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Application Information

(Continued)

LAYOUT, GROUND LOOPS AND STABILITY

The LM4733 is designed to be stable when operated at a
closed-loop gain of 10 or greater, but as with any other
high-current amplifier, the LM4733 can be made to oscillate
under certain conditions. These oscillations usually involve
printed circuit board layout or output/input coupling issues.

When designing a layout, it is important to return the load
ground, the output compensation ground, and the low level
(feedback and input) grounds to the circuit board common
ground point through separate paths. Otherwise, large cur-
rents flowing along a ground conductor will generate volt-
ages on the conductor which can effectively act as signals at
the input, resulting in high frequency oscillation or excessive
distortion. It is advisable to keep the output compensation
components and the 0.1µF supply decoupling capacitors as
close as possible to the LM4733 to reduce the effects of PCB
trace resistance and inductance. For the same reason, the
ground return paths should be as short as possible.

In general, with fast, high-current circuitry, all sorts of prob-
lems can arise from improper grounding which again can be
avoided by returning all grounds separately to a common
point. Without isolating the ground signals and returning the
grounds to a common point, ground loops may occur.

"Ground Loop" is the term used to describe situations occur-
ring in ground systems where a difference in potential exists
between two ground points. Ideally a ground is a ground, but
unfortunately, in order for this to be true, ground conductors
with zero resistance are necessary. Since real world ground
leads possess finite resistance, currents running through
them will cause finite voltage drops to exist. If two ground
return lines tie into the same path at different points there will
be a voltage drop between them. The first figure below
shows a common ground example where the positive input
ground and the load ground are returned to the supply
ground point via the same wire. The addition of the finite wire
resistance, R

, results in a voltage difference between the

two points as shown below.

20079498

The load current I

will be much larger than input bias current

, thus V

will follow the output voltage directly, i.e. in phase.

Therefore the voltage appearing at the non-inverting input is
effectively positive feedback and the circuit may oscillate. If
there was only one device to worry about then the values of
R

and R

would probably be small enough to be ignored;

however, several devices normally comprise a total system.
Any ground return of a separate device, whose output is in
phase, can feedback in a similar manner and cause insta-
bilities. Out of phase ground loops also are troublesome,
causing unexpected gain and phase errors.

The solution to most ground loop problems is to always use
a single-point ground system, although this is sometimes
impractical. The third figure above is an example of a single-
point ground system.

The single-point ground concept should be applied rigor-
ously to all components and all circuits when possible. Vio-
lations of single-point grounding are most common among
printed circuit board designs, since the circuit is surrounded
by large ground areas which invite the temptation to run a
device to the closest ground spot. As a final rule, make all
ground returns low resistance and low inductance by using
large wire and wide traces.

Occasionally, current in the output leads (which function as
antennas) can be coupled through the air to the amplifier
input, resulting in high-frequency oscillation. This normally
happens when the source impedance is high or the input
leads are long. The problem can be eliminated by placing a
small capacitor, C

, (on the order of 50pF to 500pF) across

the LM4733 input terminals. Refer to the External Compo-
nents Description section relating to component interaction
with C

REACTIVE LOADING

It is hard for most power amplifiers to drive highly capacitive
loads very effectively and normally results in oscillations or
ringing on the square wave response. If the output of the
LM4733 is connected directly to a capacitor with no series
resistance, the square wave response will exhibit ringing if
the capacitance is greater than about 0.2µF. If highly capaci-
tive loads are expected due to long speaker cables, a
method commonly employed to protect amplifiers from low
impedances at high frequencies is to couple to the load
through a 10

resistor in parallel with a 0.7µH inductor. The

inductor-resistor combination as shown in the Figure 5 iso-
lates the feedback amplifier from the load by providing high
output impedance at high frequencies thus allowing the 10

resistor to decouple the capacitive load and reduce the Q of
the series resonant circuit. The LR combination also pro-
vides low output impedance at low frequencies thus shorting
out the 10

resistor and allowing the amplifier to drive the

series RC load (large capacitive load due to long speaker
cables) directly.

INVERTING AMPLIFIER APPLICATION

The inverting amplifier configuration may be used instead of
the more common non-inverting amplifier configuration
shown in Figure 1. The inverting amplifier can have better
THD+N performance and eliminates the need for a large
capacitor (Ci) reducing cost and space requirements. The
values show in Figure 6 are only one example of an amplifier
with a gain of 20V/V (Gain = -R

). For different resistor

values, the value of R

should be eqaul to the parallel

combination of R

and Ri.

LM4733

www.national.com

Physical Dimensions

inches (millimeters)

unless otherwise noted

Non-Isolated TO-220 27-Lead Package

Order Number LM4733TA

NS Package Number TA27A

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NATIONAL'S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:

1. Life support devices or systems are devices or

systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.

2. A critical component is any component of a life

support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.

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Support Center
Email: new.feedback@nsc.com
Tel: 1-800-272-9959

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www.national.com

LM4733

Channel

30W

Audio

Power

Amplifier

with

Mute

National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.

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