Semiconductor Components Industries, LLC, 2002

September, 2002 Rev. 11

Publication Order Number:

CS51411/D

CS51411, CS51412,

CS51413, CS51414

1.5 A, 260 kHz and

520 kHz, Low Voltage Buck

Regulators with External

Bias or Synchronization

Capability

The CS5141X products are 1.5 A buck regulator ICs. These devices

are fixedfrequency operating at 260 kHz and 520 kHz. The regulators
use the V

control architecture to provide unmatched transient

response, the best overall regulation and the simplest loop
compensation for today's highspeed logic. These products
accommodate input voltages from 4.5 V to 40 V.

The CS51411 and CS51413 contain synchronization circuitry. The

CS51412 and CS51414 have the option of powering the controller
from an external 3.3 V to 6.0 V supply in order to improve efficiency,
especially in high input voltage, light load conditions.

The onchip NPN transistor is capable of providing a minimum of

1.5 A of output current, and is biased by an external "boost" capacitor
to ensure saturation, thus minimizing onchip power dissipation.
Protection circuitry includes thermal shutdown, cyclebycycle
current limiting and frequency foldback. The CS51411 and CS51413
are functionally pincompatible with the LT1375. The CS51412 and
CS51414 are functionally pincompatible with the LT1376.

Features

Architecture Provides UltraFast Transient Response, Improved

Regulation and Simplified Design

2.0% Error Amp Reference Voltage Tolerance

Switch Frequency Decrease of 4:1 in Short Circuit Conditions

Reduces Short Circuit Power Dissipation

BOOST Lead Allows "Bootstrapped" Operation to Maximize

Efficiency

Sync Function for Parallel Supply Operation or Noise Minimization

Shutdown Lead Provides PowerDown Option

µA Quiescent Current During PowerDown

Thermal Shutdown

Soft Start

PinCompatible with LT1375 and LT1376

= 1, 2, 3 or 4

= Assembly Location

WL, L

= Wafer Lot

YY, Y

= Year

WW, W

= Work Week

= Temperature Range, E or G

SO8

D SUFFIX

CASE 751

MARKING DIAGRAM

5141X

ALYWT

SYNC

SHDNB

GND

BOOST

PIN CONNECTIONS

SHDNB

BIAS

GND

BOOST

CS51412/4

CS51411/3

ORDERING INFORMATION

See detailed ordering and shipping information in the package
dimensions section on page 14 of this data sheet.

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PRODUCT SELECTION GUIDE

Part Number

Frequency

Temperature Range

Bias/Sync

CS51411E

260 kHz

°C to 85°C

Sync

CS51411G

260 kHz

°C to 70°C

Sync

CS51412E

260 kHz

°C to 85°C

Bias

CS51412G

260 kHz

°C to 70°C

Bias

CS51413E

520 kHz

°C to 85°C

Sync

CS51413G

520 kHz

°C to 70°C

Sync

CS51414E

520 kHz

°C to 85°C

Bias

CS51414G

520 kHz

°C to 70°C

Bias

SYNC

GND

SHDNB

CS51411/3

1N4148

3.3 V

µH

100

µF

100

µF

0.1

µF

Figure 1. Application Diagram, 4.5 V 16 V to 3.3 V @ 1.0 A Converter

0.1

µF

SYNC

Shutdown

4.5 V 16 V

205

127

1N5821

BOOST

MAXIMUM RATINGS*

Rating

Value

Unit

Operating Junction Temperature Range, T

40 to 150

°C

Lead Temperature Soldering:

Reflow: (SMD styles only) (Note 1)

230 peak

°C

Storage Temperature Range, T

65 to +150

°C

ESD Damage Threshold (Human Body Model)

2.0

1. 60 second maximum above 183

°C.

*The maximum package power dissipation must be observed.

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MAXIMUM RATINGS

Pin Name

Max

MIN

SOURCE

SINK

40 V

0.3 V

N/A

4.0 A

BOOST

40 V

0.3 V

N/A

100 mA

40 V

0.6 V/1.0 V, t < 50 ns

4.0 A

10 mA

7.0 V

0.3 V

1.0 mA

SHDNB

7.0 V

0.3 V

1.0 mA

SYNC

7.0 V

0.3 V

1.0 mA

BIAS

7.0 V

0.3 V

1.0 mA

50 mA

7.0 V

0.3 V

1.0 mA

GND

7.0 V

0.3 V

50 mA

1.0 mA

ELECTRICAL CHARACTERISTICS

(40

°C < T

< 125

°C (CS51411E/2E/3E/4E); 40°C < T

< 85

°C (CS51411E/2E/3E/4E);

°C < T

< 70

°C (CS51411G/2G/3G/4G), 4.5 V< V

< 40

V; unless otherwise specified.)

Characteristic

Test Conditions

Min

Typ

Max

Unit

Oscillator

Operating Frequency

CS51411/CS51412

224

260

296

kHz

Operating Frequency

CS51413/CS51414

446

520

594

kHz

Frequency Line Regulation

0.05

0.15

%/V

Maximum Duty Cycle

Frequency Foldback Threshold

0.29

0.32

0.36

PWM Comparator

Slope Compensation Voltage

CS51411/CS51412, Fix V

FB,

CS51413/CS51414

8.0

17
50

26
75

mV/

µs

mV/

µs

Minimum Output Pulse Width

CS51411/CS51412, V

to V

CS51413/CS51414, V

to V

150

300
230

ns
ns

Power Switch

Current Limit

> 0.36 V

1.6

2.3

3.0

Foldback Current

< 0.29 V

0.9

1.5

2.1

Saturation Voltage

OUT

= 1.5 A, V

BOOST

= V

+ 2.5 V

0.4

0.7

1.0

Current Limit Delay

Note 2

120

160

Error Amplifier

Internal Reference Voltage

1.244

1.270

1.296

Reference PSRR

Note 2

FB Input Bias Current

0.02

0.1

µA

Output Source Current

= 1.270 V, V

= 1.0 V

µA

Output Sink Current

= 1.270 V, V

= 2.0 V

µA

Output High Voltage

= 1.0 V

1.39

1.46

1.53

Output Low Voltage

= 2.0 V

5.0

Unity Gain Bandwidth

Note 2

500

kHz

Open Loop Amplifier Gain

Note 2

Amplifier Transconductance

Note 2

6.4

mA/V

2. Guaranteed by design, not 100% tested in production.

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ELECTRICAL CHARACTERISTICS (continued)

(40

°C < T

< 125

°C (CS51411E/2E/3E/4E); 40°C < T

< 85

°C (CS51411E/2E/3E/4E);

°C < T

< 70

°C (CS51411G/2G/3G/4G), 4.5 V< V

< 40

V; unless otherwise specified.)

Characteristic

Unit

Max

Typ

Min

Test Conditions

Sync

Sync Frequency Range

CS51411/CS51412

305

470

kHz

Sync Frequency Range

CS51413/CS51414

575

880

kHz

Sync Pin Bias Current

SYNC

= 0 V

SYNC

= 5.0 V

250

0.1

360

0.2

460

µA

Sync Threshold Voltage

1.0

1.5

1.9

Shutdown

Shutdown Threshold Voltage

1.0

1.3

1.6

Shutdown Pin Bias Current

SHDNB

= 0 V

0.14

5.00

µA

Thermal Shutdown

Overtemperature Trip Point

Note 3

175

185

195

°C

Thermal Shutdown Hysteresis

Note 3

°C

General

Quiescent Current

= 0 A

3.0

4.0

6.25

Shutdown Quiescent Current

SHDNB

= 0 V

8.0

µA

Boost Operating Current

BOOST

= 2.5 V

6.0

mA/A

Minimum Boost Voltage

Note 3

2.5

Start up Voltage

2.2

3.3

4.4

Minimum Output Current

7.0

3. Guaranteed by design, not 100% tested in production.

PACKAGE PIN DESCRIPTION

PACKAGE PIN #

PIN SYMBOL

FUNCTION

BOOST

The BOOST pin provides additional drive voltage to the onchip NPN power transistor.
The resulting decrease in switch on voltage increases efficiency.

This pin is the main power input to the IC.

This is the connection to the emitter of the onchip NPN power transistor and serves
as the switch output to the inductor. This pin may be subjected to negative voltages
during switch offtime. A catch diode is required to clamp the pin voltage in normal
operation. This node can stand 1.0 V for less than 50 ns during switch node flyback.

4 (CS51412/CS51414)

BIAS

The BIAS pin connects to the onchip power rail and allows the IC to run most of its
internal circuitry from the regulated output or another low voltage supply to improve
efficiency. The BIAS pin is left floating if this feature is not used.

5 (CS51411/CS51413)

SYNC

This pin provides the synchronization input.

5 (CS51412/CS51414)

4 (CS51411/CS51413)

SHDNB

The shutdown pin is active low and TTL compatible. The IC goes into sleep mode,
drawing less than 85

µA when the pin voltage is pulled below 1.0 V. This pin should be

left floating in normal position.

GND

Power return connection for the IC.

The FB pin provides input to the inverting input of the error amplifier. If V

is lower

than 0.29 V, the oscillator frequency is divided by four, and current limit folds back to
about 1 ampere. These features protect the IC under severe overcurrent or short
circuit conditions.

The V

pin provides a connection point to the output of the error amplifier and input to

the PWM comparator. Driving of this pin should be avoided because onchip test
circuitry becomes active whenever current exceeding 0.5 mA is forced into the IC.

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APPLICATIONS INFORMATION

THEORY OF OPERATION

Control

The CS5141X family of buck regulators provides leading

edge technology, a high level of integration and high
operating frequencies allowing the layout of a switchmode
power supply in a very small board area. These devices are
based on the proprietary V

control architecture. V

control

uses the output voltage and its ripple as the ramp signal,
providing an ease of use not generally associated with
voltage or current mode control. Improved line regulation,
load regulation and very fast transient response are also
major advantages.

Figure 3. Buck Converter with V

Control.

Buck
Controller

FFB

REF

Duty Cycle

Control

Error

Amplifier

PWM

Comparator

Oscillator

SFB

Latch

Slope

Comp

As shown in Figure 3, there are two voltage feedback

paths in V

control, namely FFB(Fast Feedback) and

SFB(Slow Feedback). In FFB path, the feedback voltage
connects directly to the PWM comparator. This feedback
path carries the ramp signal as well as the output DC voltage.
Artificial ramp derived from oscillator is added to the
feedback signal to improve stability. The other feedback
path SFB connects the feedback voltage to the error
amplifier whose output V

feeds to the other input of the

PWM comparator. In a constant frequency mode, the
oscillator signal sets the output latch and turns on the switch
S1. This starts a new switch cycle. The ramp signal,
composed of both artificial ramp and output ripple,
eventually comes across the V

voltage, and consequently

resets the latch to turn off the switch. The switch S1 will turn
on again at the beginning of the next switch cycle. In a buck
converter, the output ripple is determined by the ripple
current of the inductor L1 and the ESR (equivalent series
resistor) of the output capacitor C1.

The slope compensation signal is a fixed voltage ramp

provided by the oscillator. Adding this signal eliminates
subharmonic oscillation associated with the operation at
duty cycle greater than 50%. The artificial ramp also ensures
the proper PWM function when the output ripple voltage is
inadequate. The slope compensation signal is properly sized
to serve it purposes without sacrificing the transient
response speed.

Under load and line transient, not only the ramp signal

changes, but more significantly the DC component of the
feedback voltage varies proportionally to the output voltage.
FFB path connects both signals directly to the PWM
comparator. This allows instant modulation of the duty cycle
to counteract any output voltage deviations. The transient
response time is independent of the error amplifier
bandwidth. This eliminates the delay associated with error
amplifier and greatly improves the transient response time.
The error amplifier is used here to ensure excellent DC
accuracy.

Error Amplifier

The CS5141X has a transconductance error amplifier,

whose noninverting input is connected to an Internal
Reference Voltage generated from the onchip regulator.
The inverting input connects to the V

pin. The output of

the error amplifier is made available at the V

pin. A typical

frequency compensation requires only a 0.1

µF capacitor

connected between the V

pin and ground, as shown in

Figure 1. This capacitor and error amplifier's output
resistance (approximately 8.0 M

) create a low frequency

pole to limit the bandwidth. Since V

control does not

require a high bandwidth error amplifier, the frequency
compensation is greatly simplified.

The V

pin is clamped below Output High Voltage. This

allows the regulator to recover quickly from over current or
short circuit conditions.

Oscillator and Sync Feature (CS51411 and CS51413 only)

The onchip oscillator is trimmed at the factory and

requires no external components for frequency control. The
high switching frequency allows smaller external
components to be used, resulting in a board area and cost
savings. The tight frequency tolerance simplifies magnetic
components selection. The switching frequency is reduced
to 25% of the nominal value when the V

pin voltage is

below Frequency Foldback Threshold. In short circuit or
overload conditions, this reduces the power dissipation of
the IC and external components.

An external clock signal can sync CS51411/CS51414 to

a higher frequency. The rising edge of the sync pulse turns
on the power switch to start a new switching cycle, as shown
in Figure 4. There is approximately 0.5

µs delay between the

rising edge of the sync pulse and rising edge of the V

voltage. The sync threshold is TTL logic compatible, and
duty cycle of the sync pulses can vary from 10% to 90%. The

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frequency foldback feature is disabled during the sync
mode.

Figure 4. A CS51411 Buck Regulator is Synced by an

External 350 kHz Pulse Signal

Power Switch and Current Limit

The collector of the builtin NPN power switch is

connected to the V

pin, and the emitter to the V

pin.

When the switch turns on, the V

voltage is equal to the

minus switch Saturation Voltage. In the buck regulator,

the V

voltage swings to one diode drop below ground

when the power switch turns off, and the inductor current is
commutated to the catch diode. Due to the presence of high
pulsed current, the traces connecting the V

pin, inductor

and diode should be kept as short as possible to minimize the
noise and radiation. For the same reason, the input capacitor
should be placed close to the V

pin and the anode of the

diode.

The saturation voltage of the power switch is dependent

on the switching current, as shown in Figure 5.

Figure 5. The Saturation Voltage of the Power Switch

Increases with the Conducting Current

0.5

1.0

1.5

Switching Current (A)

(V)

0.1

0.2

0.3

0.4

0.5

0.6

0.7

Members of the CS5141X family contain pulsebypulse

current limiting to protect the power switch and external

components. When the peak of the switching current reaches
the Current Limit, the power switch turns off after the
Current Limit Delay. The switch will not turn on until the
next switching cycle. The current limit threshold is
independent of switching duty cycle. The maximum load
current, given by the following formula under continuous
conduction mode, is less than the Current Limit due to the
ripple current.

IO(MAX) + ILIM *

VO(VIN * VO)

2(L)(VIN)(fs)

where:

= switching frequency,

LIM

= current limit threshold,

= output voltage,

= input voltage,

L = inductor value.
When the regulator runs under current limit, the

subharmonic oscillation may cause low frequency
oscillation, as shown in Figure 6. Similar to current mode
control, this oscillation occurs at the duty cycle greater than
50% and can be alleviated by using a larger inductor value.
The current limit threshold is reduced to Foldback Current
when the FB pin falls below Foldback Threshold. This
feature protects the IC and external components under the
power up or overload conditions.

Figure 6. The Regulator in Current Limit

BOOST Pin

The BOOST pin provides base driving current for the

power switch. A voltage higher than V

provides required

headroom to turn on the power switch. This in turn reduces
IC power dissipation and improves overall system
efficiency. The BOOST pin can be connected to an external
booststrapping circuit which typically uses a 0.1

µF capacitor

and a 1N914 or 1N4148 diode, as shown in Figure 1. When the
power switch is turned on, the voltage on the BOOST pin is
equal to

VBOOST + VIN ) VO * VF

where:

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= diode forward voltage.

The anode of the diode can be connected to any DC voltage

other than the regulated output voltage. However, the
maximum voltage on the BOOST pin shall not exceed 40 V.

As shown in Figure 7, the BOOST pin current includes a

constant 7.0 mA predriver current and base current
proportional to switch conducting current. A detailed
discussion of this current is conducted in Thermal
Consideration section. A 0.1

µF capacitor is usually

adequate for maintaining the Boost pin voltage during the on
time.

Figure 7. The Boost Pin Current Includes 7.0 mA

PreDriver Current and Base Current when the

Switch is Turned On. The Beta Decline of the

Power Switch Further Increases the Base

Current at High Switching Current

0.5

1.0

1.5

Switching Current (A)

Boost Pin Current (mA)

BIAS Pin (CS51412 and CS51414 Only)

The BIAS pin allows a secondary power supply to bias the

control circuitry of the IC. The BIAS pin voltage should be
between 3.3 V and 6.0 V. If the BIAS pin voltage falls below
that range, use a diode to prevent current drain from the
BIAS pin. Powering the IC with a voltage lower than the
regulator's input voltage reduces the IC power dissipation
and improves energy transfer efficiency.

Shutdown

The internal power switch will not turn on until the V

pin rises above the Start Up Voltage. This ensures no
switching until adequate supply voltage is provided to the
IC.

The IC enters a sleep mode when the SHDNB pin is pulled

below Shutdown Threshold Voltage. In the sleep mode, the
power switch keeps open and the supply current reduces to
Shutdown Quiescent Current. This pin has internal pullup
current. So when this pin is not used, leave the SHDNB pin
open.

StartUp

During power up, the regulator tends to quickly charge up

the output capacitors to reach voltage regulation. This gives
rise to an excessive inrush current which can be detrimental

to the inductor, IC and catch diode. In V

control , the

compensation capacitor provides Soft Start with no need for
extra pin or circuitry. During the power up, the Output
Source Current of the error amplifier charges the
compensation capacitor which forces V

pin and thus output

voltage ramp up gradually. The Soft Start duration can be
calculated by

TSS +

VC CCOMP

ISOURCE

where:

= V

pin steadystate voltage, which is approximately

equal to error amplifier's reference voltage.

COMP

= Compensation capacitor connected to the V

SOURCE

= Output Source Current of the error amplifier.

Using a 0.1

µF C

COMP

, the calculation shows a T

over

5.0 ms which is adequate to avoid any current stresses.
Figure 8 shows the gradual rise of the V

, V

and envelope

of the V

during power up. There is no voltage overshoot

after the output voltage reaches the regulation. If the supply
voltage rises slower than the V

pin, output voltage may

overshoot.

Figure 8. The Power Up Transition of CS5141X

Regulator

Short Circuit

When the V

pin voltage drops below Foldback

Threshold, the regulator reduces the peak current limit by
40% and switching frequency to 1/4 of the nominal
frequency. These features are designed to protect the IC and
external components during over load or short circuit
conditions. In those conditions, peak switching current is
clamped to the current limit threshold. The reduced
switching frequency significantly increases the ripple
current, and thus lowers the DC current. The short circuit can
cause the minimum duty cycle to be limited by Minimum
Output Pulse Width. The foldback frequency reduces the
minimum duty cycle by extending the switching cycle. This
protects the IC from overheating, and also limits the power
that can be transferred to the output. The current limit
foldback effectively reduces the current stress on the

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inductor and diode. When the output is shorted, the DC
current of the inductor and diode can approach the current
limit threshold. Therefore, reducing the current limit by 40%
can result in an equal percentage drop of the inductor and
diode current. The short circuit waveforms are captured in
Figure 9, and the benefit of the foldback frequency and
current limit is selfevident.

Figure 9. In Short Circuit, the Foldback Current and

Foldback Frequency Limit the Switching Current to

Protect the IC, Inductor and Catch Diode

Thermal Considerations

A calculation of the power dissipation of the IC is always

necessary prior to the adoption of the regulator. The current
drawn by the IC includes quiescent current, predriver
current, and power switch base current. The quiescent
current drives the low power circuits in the IC, which
include comparators, error amplifier and other logic blocks.
Therefore, this current is independent of the switching
current and generates power equal to

WQ + VIN IQ

where:

= quiescent current.

The predriver current is used to turn on/off the power

switch and is approximately equal to 12 mA in worst case.
During steady state operation, the IC draws this current from
the Boost pin when the power switch is on and then receives
it from the V

pin when the switch is off. The predriver

current always returns to the V

pin. Since the predriver

current goes out to the regulator's output even when the
power switch is turned off, a minimum load is required to
prevent overvoltage in light load conditions. If the Boost pin
voltage is equal to V

+ V

when the switch is on, the power

dissipation due to predriver current can be calculated by

WDRV + 12 mA (VIN * VO )

VO2

VIN

)

The base current of a bipolar transistor is equal to collector

current divided by beta of the device. Beta of 60 is used here
to estimate the base current. The Boost pin provides the base

current when the transistor needs to be on. The power
dissipated by the IC due to this current is

WBASE +

VO2

VIN

where:

= DC switching current.

When the power switch turns on, the saturation voltage

and conduction current contribute to the power loss of a
nonideal switch. The power loss can be quantified as

WSAT +

VIN

IS VSAT

where:

SAT

= saturation voltage of the power switch which is

shown in Figure 5.

The switching loss occurs when the switch experiences

both high current and voltage during each switch transition.
This regulator has a 30 ns turnoff time and associated
power loss is equal to

WS +

IS VIN

20 ns fS

The turnon time is much shorter and thus turnon loss is

not considered here.

The total power dissipated by the IC is sum of all the above

WIC + WQ ) WDRV ) WBASE ) WSAT ) WS

The IC junction temperature can be calculated from the

ambient temperature, IC power dissipation and thermal
resistance of the package. The equation is shown as follows,

TJ + WIC R

qJA ) TA

The maximum IC junction temperature shall not exceed

125

°C to guarantee proper operation and avoid any damages

to the IC.

Minimum Load Requirement

As pointed out in the previous section, a minimum load is

required for this regulator due to the predriver current
feeding the output. Placing a resistor equal to V

divided by

12 mA should prevent any voltage overshoot at light load
conditions. Alternatively, the feedback resistors can be
valued properly to consume 12 mA current.

COMPONENT SELECTION

Input Capacitor

In a buck converter, the input capacitor witnesses pulsed

current with an amplitude equal to the load current. This
pulsed current and the ESR of the input capacitors determine
the V

ripple voltage, which is shown in Figure 10. For V

ripple, low ESR is a critical requirement for the input
capacitor selection. The pulsed input current possesses a
significant AC component, which is absorbed by the input
capacitors. The RMS current of the input capacitor can be
calculated using:

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IRMS + IO D(1 * D)

where:

D = switching duty cycle which is equal to V

= load current.

Figure 10. Input Voltage Ripple in a Buck Converter

To calculate the RMS current, multiply the load current

with the constant given by Figure 11 at each duty cycle. It is
a common practice to select the input capacitor with an RMS
current rating more than half the maximum load current. If
multiple capacitors are paralleled, the RMS current for each
capacitor should be the total current divided by the number
of capacitors.

Figure 11. Input Capacitor RMS Current can be

Calculated by Multiplying Y Value with Maximum Load

Current at any Duty Cycle

0.2

0.4

1.0

Duty Cycle

0.1

0.3

0.4

0.5

0.6

0.2

0.6

0.8

RMS

(XI

)

Selecting the capacitor type is determined by each

design's constraint and emphasis. The aluminum
electrolytic capacitors are widely available at lowest cost.
Their ESR and ESL (equivalent series inductor) are
relatively high. Multiple capacitors are usually paralleled to
achieve lower ESR. In addition, electrolytic capacitors
usually need to be paralleled with a ceramic capacitor for
filtering high frequency noises. The OSCON are solid
aluminum electrolytic capacitors, and therefore has a much
lower ESR. Recently, the price of the OSCON capacitors
has dropped significantly so that it is now feasible to use
them for some low cost designs. Electrolytic capacitors are
physically large, and not used in applications where the size,
and especially height is the major concern.

Ceramic capacitors are now available in values over 10

µF.

Since the ceramic capacitor has low ESR and ESL, a single
ceramic capacitor can be adequate for both low frequency
and high frequency noises. The disadvantage of ceramic
capacitors are their high cost. Solid tantalum capacitors can
have low ESR and small size. However, the reliability of the
tantalum capacitor is always a concern in the application
where the capacitor may experience surge current.

Output Capacitor

In a buck converter, the requirements on the output

capacitor are not as critical as those on the input capacitor.
The current to the output capacitor comes from the inductor
and thus is triangular. In most applications, this makes the
RMS ripple current not an issue in selecting output
capacitors.

The output ripple voltage is the sum of a triangular wave

caused by ripple current flowing through ESR, and a square
wave due to ESL. Capacitive reactance is assumed to be
small compared to ESR and ESL. The peak to peak ripple
current of the inductor is:

* P +

VO(VIN * VO)

(VIN)(L)(fS)

RIPPLE(ESR)

, the output ripple due to the ESR, is equal

to the product of I

and ESR. The voltage developed

across the ESL is proportional to the di/dt of the output
capacitor. It is realized that the di/dt of the output capacitor
is the same as the di/dt of the inductor current. Therefore,
when the switch turns on, the di/dt is equal to (V

)/L,

and it becomes V

/L when the switch turns off. The total

ripple voltage induced by ESL can then be derived from

VRIPPLE(ESL) + ESL(

VIN

)

) ESL(

VIN * VO

)

+ ESL(

VIN

)

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The total output ripple is the sum of the V

RIPPLE(ESR)

and

RIPPLE(ESR)

Figure 12. The Output Voltage Ripple Using Two 10

Ceramic Capacitors in Parallel

Figure 13. The Output Voltage Ripple Using One 100

POSCAP Capacitor

Figure 14. The Output Voltage Ripple Using

One 100

mF OSCON

Figure 15. The Output Voltage Ripple Using

One 100

mF Tantalum Capacitor

Figure 12 to Figure 15 show the output ripple of a 5.0 V

to 3.3 V/500 mA regulator using 22

µH inductor and various

capacitor types. At the switching frequency, the low ESR
and ESL make the ceramic capacitors behave capacitively
as shown in Figure 12. Additional paralleled ceramic
capacitors will further reduce the ripple voltage, but
inevitably increase the cost. "POSCAP", manufactured by
SANYO, is a solid electrolytic capacitor. The anode is
sintered tantalum and the cathode is a highly conductive
polymerized organic semiconductor. TPC series, featuring
low ESR and low profile, is used in the measurement of
Figure 13. It is shown that POSCAP presents a good balance
of capacitance and ESR, compared with a ceramic capacitor.
In this application, the low ESR generates less than 5.0 mV
of ripple and the ESL is almost unnoticeable. The ESL of the
throughhole OSCON capacitor give rise to the inductive
impedance. It is evident from Figure 14 which shows the
step rise of the output ripple on the switch turnon and large
spike on the switch turnoff. The ESL prevents the output
capacitor from quickly charging up the parasitic capacitor of
the inductor when the switch node is pulled below ground
through the catch diode conduction. This results in the spike
associated with the falling edge of the switch node. The D
package tantalum capacitor used in Figure 15 has the same
footprint as the POSCAP, but doubles the height. The ESR
of the tantalum capacitor is apparently higher than the
POSCAP. The electrolytic and tantalum capacitors provide
a lowcost solution with compromised performance. The
reliability of the tantalum capacitor is not a serious concern
for output filtering because the output capacitor is usually
free of surge current and voltage.

Diode Selection

The diode in the buck converter provides the inductor

current path when the power switch turns off. The peak
reverse voltage is equal to the maximum input voltage. The
peak conducting current is clamped by the current limit of
the IC. The average current can be calculated from:

CS51411, CS51412, CS51413, CS51414

http://onsemi.com

ID(AVG) +

IO(VIN * VO)

VIN

The worse case of the diode average current occurs during

maximum load current and maximum input voltage. For the
diode to survive the short circuit condition, the current rating
of the diode should be equal to the Foldback Current Limit.
See Table 1 for schottky diodes from ON Semiconductor
which are suggested for CS5141X regulator.

Inductor Selection

When choosing inductors, one might have to consider

maximum load current, core and copper losses, component
height, output ripple, EMI, saturation and cost. Lower
inductor values are chosen to reduce the physical size of the
inductor. Higher value cuts down the ripple current, core
losses and allows more output current. For most
applications, the inductor value falls in the range between
2.2

µH and 22 µH. The saturation current ratings of the

inductor shall not exceed the I

L(PK)

, calculated according to

IL(PK) + IO )

VO(VIN * VO)

2(fS)(L)(VIN)

The DC current through the inductor is equal to the load

current. The worse case occurs during maximum load
current. Check the vendor's spec to adjust the inductor value
under current loading. Inductors can lose over 50% of
inductance when it nears saturation.

The core materials have a significant effect on inductor

performance. The ferrite core has benefits of small physical
size, and very low power dissipation. But be careful not to
operate these inductors too far beyond their maximum
ratings for peak current, as this will saturate the core.
Powered Iron cores are low cost and have a more gradual
saturation curve. The cores with an open magnetic path, such
as rod or barrel, tend to generate high magnetic field
radiation. However, they are usually cheap and small. The
cores providing a close magnetic loop, such as potcore and
toroid, generate low electromagnetic interference (EMI).

There are many magnetic component vendors providing

standard product lines suitable for CS5141X. Table 2 lists
three vendors, their products and contact information.

Table 1.

Part Number

BREAKDOWN

(V)

AVERAGE

(A)

(F)

(V) @ I

AVERAGE

Package

1N5817

1.0

0.45

Axial Lead

1N5818

1.0

0.55

Axial Lead

1N5819

1.0

0.6

Axial Lead

MBR0520

0.5

0.385

SOD123

MBR0530

0.5

0.43

SOD123

MBR0540

0.5

0.53

SOD123

MBRS120

1.0

0.55

SMB

MBRS130

1.0

0.395

SMB

MBRS140

1.0

0.6

SMB

Table 2.

Vendor

Product Family

Web Site

Telephone

Coiltronics

UNIPac1/2: SMT, barrel

THINPAC: SMT, toroid, low profile

CTX: Leaded, toroid

www.coiltronics.com

(516) 2417876

Coilcraft

DO1608: SMT, barrel

DS/DT 1608: SMT, barrel, magnetically shielded

DO3316: SMT, barrel

DS/DT 3316: SMT, barrel, magnetically shielded

DO3308: SMT, barrel, low profile

www.coilcraft.com

(800) 3222645

Pulse

www.pulseeng.com

(619) 6748100

CS51411, CS51412, CS51413, CS51414

http://onsemi.com

PACKAGE DIMENSIONS

SO8

D SUFFIX

CASE 75107

ISSUE V

SEATING

PLANE

X 45

NOTES:

1. DIMENSIONING AND TOLERANCING PER ANSI

Y14.5M, 1982.

2. CONTROLLING DIMENSION: MILLIMETER.

3. DIMENSION A AND B DO NOT INCLUDE MOLD

PROTRUSION.

4. MAXIMUM MOLD PROTRUSION 0.15 (0.006) PER

SIDE.

5. DIMENSION D DOES NOT INCLUDE DAMBAR

PROTRUSION. ALLOWABLE DAMBAR

PROTRUSION SHALL BE 0.127 (0.005) TOTAL IN

EXCESS OF THE D DIMENSION AT MAXIMUM

MATERIAL CONDITION.

0.10 (0.004)

X

Y

0.25 (0.010)

Z

0.25 (0.010)

DIM

MIN

MAX

MIN

MAX

INCHES

4.80

5.00

0.189

0.197

MILLIMETERS

3.80

4.00

0.150

0.157

1.35

1.75

0.053

0.069

0.33

0.51

0.013

0.020

1.27 BSC

0.050 BSC

0.10

0.25

0.004

0.010

0.19

0.25

0.007

0.010

0.40

1.27

0.0160.050

0.25

0.50

0.010

0.020

5.80

6.20

0.228

0.244

PACKAGE THERMAL DATA

Parameter

SO8

Unit

Typical

°C/W

Typical

165

°C/W

CS51411, CS51412, CS51413, CS51414

http://onsemi.com

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changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any
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liability, including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or
specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be
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JAPAN: ON Semiconductor, Japan Customer Focus Center

291 Kamimeguro, Meguroku, Tokyo, Japan 1530051
Phone: 81357733850
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For additional information, please contact your local
Sales Representative.

CS51411/D

is a trademark of Switch Power, Inc.

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