Dual Wideband, Current-Feedback

OPERATIONAL AMPLIFIER With Disable

DESCRIPTION

The OPA2691 sets a new level of performance for broadband dual
current-feedback op amps. Operating on a very low 5.1mA/ch
supply current, the OPA2691 offers a slew rate and output power
normally associated with a much higher supply current. A new
output stage architecture delivers a high output current with minimal
voltage headroom and crossover distortion. This gives exceptional
single-supply operation. Using a single +5V supply, the OPA2691
can deliver a 1V to 4V output swing with over 150mA drive current
and 190MHz bandwidth. This combination of features makes the
OPA2691 an ideal RGB line driver or single-supply Analog-to-Digital
Converter (ADC) input driver.

FEATURES

FLEXIBLE SUPPLY RANGE:

+5V to +12V Single Supply

2.5V to

6V Dual Supply

WIDEBAND +5V OPERATION: 190MHz (G = +2)

UNITY-GAIN STABLE: 280MHz (G = 1)

HIGH OUTPUT CURRENT: 190mA

OUTPUT VOLTAGE SWING:

4.0V

HIGH SLEW RATE: 2100V/

LOW SUPPLY CURRENT: 5.1mA/ch

LOW DISABLED CURRENT: 150

A/ch

APPLICATIONS

xDSL LINE DRIVER / RECEIVER

MATCHED I/Q CHANNEL AMPLIFIER

BROADBAND VIDEO BUFFERS

HIGH-SPEED IMAGING CHANNELS

PORTABLE INSTRUMENTS

DIFFERENTIAL ADC DRIVERS

ACTIVE FILTERS

WIDEBAND INVERTING SUMMING

The OPA2691's low 5.1mA/ch supply current is precisely trimmed at
25

C. This trim, along with low drift over temperature, ensures lower

maximum supply current than competing products. System power
may be further reduced by using the optional disable control pin
(SO-14 only). Leaving this disable pin open, or holding it HIGH,
gives normal operation. If pulled LOW, the OPA2691 supply current
drops to less than 150

A/ch while the output goes into a high

impedance state. This feature may be used for power savings.

Single-Supply ADSL Upstream Driver

100

12.4

100

2Vp-p

324

1/2

OPA2691

1/2

OPA2691

+12V

1:2

15Vp-p

12.4

+6.0V

OPA2691 RELATED PRODUCTS

SINGLES

DUALS

TRIPLES

Voltage-Feedback

OPA690

OPA2690

OPA3690

Current-Feedback

OPA691

OPA2681

OPA3691

Fixed Gain

OPA692

OPA3692

OPA2

691

OPA2

691

OPA2691

SBOS224A DECEMBER 2001 REVISED OCTOBER 2002

www.ti.com

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.

Frequency (MHz)

0.1

100 200

SINGLE-SUPPLY ADSL UPSTREAM DRIVER

SMALL-SIGNAL FREQUENCY RESPONSE

Gain (dB)

OPA2691

SBOS224A

www.ti.com

SPECIFIED

PACKAGE

TEMPERATURE

PACKAGE

ORDERING

TRANSPORT

PRODUCT

PACKAGE-LEAD

DESIGNATOR

(1)

RANGE

MARKING

NUMBER

MEDIA, QUANTITY

OPA2691

SO-8

C to +85

OPA2691

OPA2691ID

Rails, 100

OPA2691IDR

Tape and Reel, 2500

OPA2691

SO-14

C to +85

OPA2691

OPA2691I-14D

Rails, 58

OPA2691I-14DR

Tape and Reel, 2500

NOTE: (1) For the most current specifications and package information, refer to our web site at www.ti.com.

NC = No Connection

In A

+In A

DIS A

DIS B

+In B

In B

Out A

Out B

ABSOLUTE MAXIMUM RATINGS

(1)

Power Supply ...............................................................................

6.5V

Internal Power Dissipation

(2)

............................ See Thermal Information

Differential Input Voltage ..................................................................

1.2V

Input Voltage Range ...........................................................................

Storage Temperature Range: D, 14D ............................ 40

C to +125

Lead Temperature (soldering, 10s) .............................................. +300

Junction Temperature (T

) ........................................................... +175

ESD Performance:

HBM .............................................................................................. 2000V
CDM .............................................................................................. 1500V

NOTES:: (1) Stresses above those listed under "Absolute Maximum Ratings"
may cause permanent damage to the device. Exposure to absolute maximum
conditions for extended periods may affect device reliability. (2) Packages
must be derated based on specified

. Maximum T

must be observed.

PIN CONFIGURATIONS

Top View

SO-8

SO-14

Out B

In B

+In B

Out A

In A

+In A

PACKAGE/ORDERING INFORMATION

ELECTROSTATIC
DISCHARGE SENSITIVITY

This integrated circuit can be damaged by ESD. Texas Instru-
ments recommends that all integrated circuits be handled with
appropriate precautions. Failure to observe proper handling
and installation procedures can cause damage.

ESD damage can range from subtle performance degradation
to complete device failure. Precision integrated circuits may be
more susceptible to damage because very small parametric
changes could cause the device not to meet its published
specifications.

OPA2691

SBOS224A

www.ti.com

ELECTRICAL CHARACTERISTICS: V

Boldface limits are tested at +25

= 402

, R

= 100

, and G = +2

(see Figure 1 for AC performance only), unless otherwise noted.

OPA2691ID, I-14D

TYP

MIN/MAX OVER TEMPERATURE

C to

40

C to

MIN/

TEST

PARAMETER

CONDITIONS

+25

(1)

(2)

+85

(2)

UNITS

MAX

LEVEL

(3)

AC PERFORMANCE (see Figure 1)
Small-Signal Bandwidth (V

= 0.5Vp-p)

G = +1, R

= 453

280

MHz

typ

G = +2, R

= 402

225

200

190

180

MHz

min

G = +5, R

= 261

210

MHz

typ

G = +10, R

= 180

200

MHz

typ

Bandwidth for 0.1dB Gain Flatness

G = +2, V

= 0.5Vp-p

MHz

min

Peaking at a Gain of +1

= 453, V

= 0.5Vp-p

0.2

1.5

max

Large-Signal Bandwidth

G = +2, V

= 5Vp-p

200

MHz

typ

Slew Rate

G = +2, 4V Step

2100

1400

1375

1350

min

Rise-and-Fall Time

G = +2, V

= 0.5V Step

1.6

typ

G = +2, 5V Step

1.9

typ

Settling Time to 0.02%

G = +2, V

= 2V Step

typ

0.1%

G = +2, V

= 2V Step

typ

Harmonic Distortion

G = +2, f = 5MHz, V

= 2Vp-p

2nd-Harmonic

= 100

dBc

max

500

dBc

max

3rd-Harmonic

= 100

dBc

max

500

dBc

max

Input Voltage Noise

f > 1MHz

1.7

2.5

2.9

3.1

nV/

max

Noninverting Input Current Noise

f > 1MHz

pA/

max

Inverting Input Current Noise

f > 1MHz

pA/

max

Differential Gain

G = +2, NTSC, V

= 1.4Vp, R

= 150

0.07

typ

= 37.5

0.17

typ

Differential Phase

G = +2, NTSC, V

= 1.4Vp, R

= 150

0.02

deg

typ

= 37.5

0.07

deg

typ

Channel-to-Channel Crosstalk

f = 5MHz

dBc

typ

DC PERFORMANCE

(4)

Open-Loop Transimpedance Gain (Z

)

= 0V, R

= 100

225

125

110

100

min

Input Offset Voltage

= 0V

0.8

3.7

4.3

max

Average Offset Voltage Drift

= 0V

max

Noninverting Input Bias Current

= 0V

+15

+35

+43

+45

max

Average Noninverting Input Bias Current Drift

= 0V

300

nA/

max

Inverting Input Bias Current

= 0V

max

Average Inverting Input Bias Current Drift

= 0V

200

max

INPUT
Common-Mode Input Range (CMIR)

(5)

3.5

3.4

3.3

3.2

min

Common-Mode Rejection (CMRR)

= 0V

min

Noninverting Input Impedance

100 || 2

|| pF

typ

Inverting Input Resistance (R

)

Open-Loop

typ

OUTPUT
Voltage Output Swing

No Load

4.0

3.8

3.7

3.6

min

100

Load

3.9

3.7

3.6

3.3

min

Current Output, Sourcing

= 0

+190

+160

+140

+100

min

Current Output, Sinking

= 0

190

160

140

100

min

Short-Circuit Current

250

typ

Closed-Loop Output Impedance

G = +2, f = 100kHz

0.03

typ

DISABLE (Disabled LOW) (SO-14 only)
Power-Down Supply Current (+V

)

DIS

= 0, Both Channels

300

600

700

800

max

Disable Time

= 1V

400

typ

Enable Time

= 1V

typ

Off Isolation

G = +2, 5MHz

typ

Output Capacitance in Disable

typ

Turn-On Glitch

G = +2, R

= 150

, V

= 0

typ

Turn-Off Glitch

G = +2, R

= 150

, V

= 0

typ

Enable Voltage

3.3

3.5

3.6

3.7

min

Disable Voltage

1.8

1.7

1.6

1.5

max

Control Pin Input Bias Current (DIS )

DIS

= 0, Each Channel

130

150

160

max

POWER SUPPLY
Specified Operating Voltage

typ

Maximum Operating Voltage Range

max

Max Quiescent Current

5V, Both Channels

10.2

10.6

11.2

11.5

max

Min Quiescent Current

5V, Both Channels

10.2

9.8

9.2

8.9

min

Power-Supply Rejection Ratio (PSRR)

Input Referred

min

TEMPERATURE RANGE
Specification: D, 14D

40 to +85

typ

Thermal Resistance,

Junction-to-Ambient

SO-8

125

C/W

typ

14D

SO-14

100

C/W

typ

NOTES: (1) Junction temperature = ambient for +25

C specifications. (2) Junction temperature = ambient at low temperature limit: junction temperature = ambient +15

at high temperature limit for over temperature specifications. (3) Test Levels: (A) 100% tested at +25

C. Over-temperature limits by characterization and simulation.

(B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. V

is the input common-mode

voltage. (5) Tested < 3dB below minimum specified CMRR at

CMIR limits.

OPA2691

SBOS224A

www.ti.com

ELECTRICAL CHARACTERISTICS: V

= +5V

Boldface limits are tested at +25

= 453

, R

= 100

to V

/2, and G = +2

(see Figure 1 for AC performance only), unless otherwise noted.

OPA2691ID, I-14D

TYP

MIN / MAX OVER TEMPERATURE

C to

40

C to

MIN/

TEST

PARAMETER

CONDITIONS

+25

(1)

(2)

+85

(2)

UNITS

MAX

LEVEL

(3)

AC PERFORMANCE (see Figure 2)
Small-Signal Bandwidth (V

= 0.5Vp-p)

G = +1, R

= 499

210

MHz

typ

G = +2, R

= 453

190

168

160

140

MHz

min

G = +5, R

= 340

180

MHz

typ

G = +10, R

= 180

155

MHz

typ

Bandwidth for 0.1dB Gain Flatness

G = +2, V

< 0.5Vp-p

MHz

min

Peaking at a Gain of +1

= 649

, V

< 0.5Vp-p

0.2

2.5

3.0

max

Large-Signal Bandwidth

G = +2, V

= 2Vp-p

210

MHz

typ

Slew Rate

G = +2, 2V Step

850

600

575

550

min

Rise-and-Fall Time

G = +2, V

= 0.5V Step

2.0

typ

G = +2, V

= 2V Step

2.3

typ

Settling Time to 0.02%

G = +2, V

= 2V Step

typ

0.1%

G = +2, V

= 2V Step

typ

Harmonic Distortion

G = +2, f = 5MHz, V

= 2Vp-p

2nd-Harmonic

= 100

to V

/ 2

dBc

max

500

to V

dBc

max

3rd-Harmonic

= 100

to V

/ 2

dBc

max

500

to V

dBc

max

Input Voltage Noise

f > 1MHz

1.7

2.5

2.9

3.1

nV/

max

Noninverting Input Current Noise

f > 1MHz

pA/

max

Inverting Input Current Noise

f > 1MHz

pA/

max

DC PERFORMANCE

(4)

Open-Loop Transimpedance Gain (Z

)

= V

/2, R

= 100

to V

200

100

min

Input Offset Voltage

= 2.5V

0.8

3.5

4.1

4.8

max

Average Offset Voltage Drift

= 2.5V

max

Noninverting Input Bias Current

= 2.5V

+20

+40

+48

+56

max

Average Noninverting Input Bias Current Drift

= 2.5V

250

nA/

max

Inverting Input Bias Current

= 2.5V

max

Average Inverting Input Bias Current Drift

= 2.5V

112

250

nA /

max

INPUT
Least Positive Input Voltage

(5)

1.5

1.6

1.7

1.8

max

Most Positive Input Voltage

(5)

3.5

3.4

3.3

3.2

min

Common-Mode Rejection (CMRR)

= 2.5V

min

Noninverting Input Impedance

100 || 2

|| pF

typ

Inverting Input Resistance (R

)

Open-Loop

typ

OUTPUT
Most Positive Output Voltage

No Load

3.8

3.7

3.5

min

= 100

, 2.5V

3.9

3.7

3.6

3.4

min

Least Positive Output Voltage

No Load

1.2

1.3

1.5

max

= 100

, 2.5V

1.1

1.3

1.4

1.6

max

Current Output, Sourcing

= V

+160

+120

+100

+80

min

Current Output, Sinking

= V

160

120

100

min

Closed-Loop Output Impedance

G = +2, f = 100kHz

0.03

typ

DISABLE (Disabled LOW) (SO-14 only)
Power-Down Supply Current (+V

)

DIS

= 0, Both Channels

300

600

700

800

max

Off Isolation

G = +2, 5MHz

typ

Output Capacitance in Disable

typ

Turn-On Glitch

G = +2, R

= 150

, V

= V

typ

Turn-Off Glitch

G = +2, R

= 150

, V

= V

typ

Enable Voltage

3.3

3.5

3.6

3.7

min

Disable Voltage

1.8

1.7

1.6

1.5

max

Control Pin Input Bias Current (DIS )

DIS

= 0, Each Channel

130

150

160

typ

POWER SUPPLY
Specified Single-Supply Operating Voltage

typ

Maximum Single-Supply Operating Voltage

max

Max Quiescent Current

= +5V, Both Channels

9.6

10.4

max

Min Quiescent Current

= +5V, Both Channels

8.2

8.0

7.8

min

Power-Supply Rejection Ratio (+PSRR)

Input Referred

typ

TEMPERATURE RANGE
Specification: D, 14D

40 to +85

typ

Thermal Resistance,

SO-8

125

C/W

typ

14D

SO-14

100

C/W

typ

NOTES: (1) Junction temperature = ambient for +25

C specifications. (2) Junction temperature = ambient at low temperature limit: junction temperature = ambient +15

at high temperature limit for over temperature specifications. (3) Test Levels: (A) 100% tested at +25

C. Over-temperature limits by characterization and simulation.

(B) Limits set by characterization and simulation. (C) Typical value only for information. (4) Current is considered positive out of node. V

is the input common-mode

voltage. (5) Tested < 3dB below minimum specified CMRR at

CMIR limits.

OPA2691

SBOS224A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

G = +2, R

= 402

, R

= 100

, unless otherwise noted (see Figure 1 for AC performance only).

100

HARMONIC DISTORTION vs LOAD RESISTANCE

Load Resistance (

)

100

1000

Harmonic Distortion (dBc)

= 2Vp-p

f = 5MHz

3rd-Harmonic

2nd-Harmonic

HARMONIC DISTORTION vs SUPPLY VOLTAGE

Supply Voltage (V)

2.5

3.5

4.5

5.5

Harmonic Distortion (dBc)

= 2Vp-p

= 100

f = 5MHz

3rd-Harmonic

2nd-Harmonic

100

HARMONIC DISTORTION vs FREQUENCY

Frequency (MHz)

0.1

Harmonic Distortion (dBc)

dBc = dB Below Carrier

= 2Vp-p

= 100

3rd-Harmonic

2nd-Harmonic

HARMONIC DISTORTION vs OUTPUT VOLTAGE

Output Voltage Swing (Vp-p)

0.1

Harmonic Distortion (dBc)

= 100

f = 5MHz

3rd-Harmonic

2nd-Harmonic

HARMONIC DISTORTION vs NONINVERTING GAIN

Gain (V/V)

Harmonic Distortion (dBc)

= 2Vp-p

= 100

f = 5MHz

2nd-Harmonic

3rd-Harmonic

HARMONIC DISTORTION vs INVERTING GAIN

Inverting Gain (V/V)

Harmonic Distortion (dBc)

= 2Vp-p

= 100

f = 5MHz

= 402

3rd-Harmonic

2nd-Harmonic

OPA2691

SBOS224A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

G = +2, R

= 402

, R

= 100

, unless otherwise noted (see Figure 1 for AC performance only).

100

INPUT VOLTAGE AND CURRENT NOISE DENSITY

Frequency (Hz)

100

10k

100k

10M

Current Noise (pA/

Hz)

Voltage Noise (nV/

Hz)

Noninverting Input Current Noise (12pA/

Hz)

Inverting Input Current Noise (15pA/

Hz)

Voltage Noise (1.7nV/

Hz)

2-TONE, 3RD-ORDER

INTERMODULATION SPURIOUS

Single-Tone Load Power (dBm)

3rd-Order Spurious Level (dBc)

dBc = dB below carriers

50MHz

20MHz

10MHz

Load Power at Matched 50

Load

RECOMMENDED R

vs CAPACITIVE LOAD

Capacitive Load (pF)

100

(

)

Frequency (25MHz/div)

250MHz

125MHz

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Normalized Gain to Capacitive Load (dB)

1/2

OPA2691

402

is optional.

= 22pF

= 10pF

= 47pF

= 100pF

2.0

1.6

1.2

0.8

0.4

LARGE-SIGNAL DISABLE/ENABLE RESPONSE

Time (200ns/div)

Output Voltage (400mV/div)

6.0

4.0

2.0

DIS

(2V/div)

DIS

Output Voltage

= +1V

DISABLED FEEDTHROUGH vs FREQUENCY

100

Frequency (MHz)

0.3

100

Feedthrough (5dB/div)

Forward

DIS

= 0

Reverse

OPA2691

SBOS224A

www.ti.com

TYPICAL CHARACTERISTICS: V

(Cont.)

G = +2, R

= 402

, R

= 100

, unless otherwise noted (see Figure 1 for AC performance only).

OUTPUT VOLTAGE AND CURRENT LIMITATIONS

(mA)

150

200

250

300

100

+100

+50

+200

+150

+250 +300

(V)

100

Load Line

Output Current Limit

1W Internal

Power Limit

Single Channel

1W Internal

Power Limit

Single Channel

Output Current Limit

CMRR AND PSRR vs FREQUENCY

Frequency (Hz)

10k

100k

10M

100M

Common-Mode Rejection Ratio (dB)

Power-Supply Rejection Ratio (dB)

+PSRR

CMRR

PSRR

0.1

0.01

CLOSED-LOOP OUTPUT IMPEDANCE

vs FREQUENCY

Frequency (Hz)

10k

100M

100k

10M

Output Impedance (

)

1/2

OPA2691

402

120

100

OPEN-LOOP TRANSIMPEDANCE GAIN/PHASE

Frequency (Hz)

10k

100k

10M

100M

Transimpedance Gain (20dB/div)

120

160

200

240

Transimpedance Phase (40

/div)

| Z

1.5

0.5

1.5

TYPICAL DC DRIFT OVER TEMPERATURE

Ambient Temperature (

100

125

Input Offset Voltage (mV)

Input Bias Currents (

Noninverting Input Bias Current (I

B+

)

Inverting Input

Bias Current (I

)

Input Offset

Voltage (V

)

250

200

150

100

SUPPLY AND OUTPUT CURRENT vs TEMPERATURE

Ambient Temperature (

100

125

Supply Current (2mA/div)

Output Current (50mA/div)

Quiescent Supply Current
Both Channels

Sourcing Output Current

Sinking Output Current

OPA2691

SBOS224A

www.ti.com

TYPICAL CHARACTERISTICS: V

= +5V

G = +2, R

= 499

, R

= 100

to +2.5V, unless otherwise noted (see Figure 2 for AC performance only).

Frequency (25MHz/div)

250MHz

125MHz

SMALL-SIGNAL FREQUENCY RESPONSE

Normalized Gain (1dB/div)

G = +2,

= 453

G = +10,

= 180

G = +5,

= 340

G = +1,

= 499

= 0.5Vp-p

7.0

6.5

6.0

5.5

5.0

4.5

4.0

3.5

3.0

2.5

2.0

Frequency (25MHz/div)

250MHz

125MHz

LARGE-SIGNAL FREQUENCY RESPONSE

Gain (0.5dB/div)

= 0.5Vp-p

= 2Vp-p

G = +2
R

= 100

to 2.5V

= 1Vp-p

2.9

2.8

2.7

2.6

2.5

2.4

2.3

2.2

2.1

SMALL-SIGNAL PULSE RESPONSE

Time (5ns/div)

Output Voltage (100mV/div)

G = +2

= 0.5Vp-p

4.1

3.7

3.3

2.9

2.5

2.1

1.7

1.3

0.9

LARGE-SIGNAL PULSE RESPONSE

Time (5ns/div)

Output Voltage (400mV/div)

G = +2

= 2Vp-p

RECOMMENDED R

vs CAPACITIVE LOAD

Capacitive Load (pF)

100

(

)

FREQUENCY RESPONSE vs CAPACITIVE LOAD

Frequency (25MHz/div)

250MHz

125MHz

Normalized Gain to Capacitive Load (dB)

= 22pF

= 10pF

= 47pF

= 100pF

1/2

OPA2691

453

57.6

806

+5V

0.1

is optional.

OPA2691

SBOS224A

www.ti.com

APPLICATIONS INFORMATION

WIDEBAND CURRENT-FEEDBACK OPERATION

The OPA2691 gives the exceptional AC performance of a
wideband current-feedback op amp with a highly linear, high-
power output stage. Requiring only 5.1mA/ch quiescent cur-
rent, the OPA2691 will swing to within 1V of either supply rail
and deliver in excess of 160mA tested at room temperature.
This low output headroom requirement, along with supply
voltage independent biasing, gives remarkable single (+5V)
supply operation. The OPA2691 will deliver greater than 200MHz
bandwidth driving a 2Vp-p output into 100

on a single +5V

supply. Previous boosted output stage amplifiers have typically
suffered from very poor crossover distortion as the output
current goes through zero. The OPA2691 achieves a compa-
rable power gain with much better linearity. The primary
advantage of a current-feedback op amp over a voltage-
feedback op amp is that AC performance (bandwidth and
distortion) is relatively independent of signal gain. For similar
AC performance with improved DC accuracy, consider the high
slew rate, unity-gain stable, voltage-feedback OPA2690.

Figure 1 shows the DC-coupled, gain of +2, dual power-
supply circuit configuration used as the basis of the

Electrical Characteristics and Typical Characteristics. For
test purposes, the input impedance is set to 50

with a

resistor to ground and the output impedance is set to 50

with a series output resistor. Voltage swings reported in the
electrical characteristics are taken directly at the input and
output pins while load powers (dBm) are defined at a matched
50

load. For the circuit of Figure 1, the total effective load

will be 100

|| 804

= 89

. The disable control line (DIS) is

typically left open (SO-14 only) to ensure normal amplifier
operation. One optional component is included in Figure 1. In
addition to the usual power-supply decoupling capacitors to
ground, a 0.01

F capacitor is included between the two

power-supply pins. In practical PC board layouts, this op-
tional added capacitor will typically improve the 2nd-har-
monic distortion performance by 3dB to 6dB.

Figure 2 shows the AC-coupled, gain of +2, single-supply
circuit configuration used as the basis of the +5V Electrical
Characteristics and Typical Characteristics. Though not a
"rail-to-rail" design, the OPA2691 requires minimal input and
output voltage headroom compared to other very wideband
current-feedback op amps. It will deliver a 3Vp-p output
swing on a single +5V supply with greater than 150MHz
bandwidth. The key requirement of broadband single-supply
operation is to maintain input and output signal swings within
the usable voltage ranges at both the input and the output.
The circuit of Figure 2 establishes an input midpoint bias
using a simple resistive divider from the +5V supply (two
806

resistors). The input signal is then AC-coupled into this

midpoint voltage bias. The input voltage can swing to within
1.5V of either supply pin, giving a 2Vp-p input signal range
centered between the supply pins. The input impedance
matching resistor (57.6

) used for testing is adjusted to give

a 50

input match when the parallel combination of the

biasing divider network is included. The gain resistor (R

) is

AC-coupled, giving the circuit a DC gain of +1--which puts
the input DC bias voltage (2.5V) on the output as well. The
feedback resistor value has been adjusted from the bipolar
supply condition to re-optimize for a flat frequency response
in +5V operation, gain of +2, operation (see the Setting
Resistor Values to Optimize Bandwidth section). Again, on a
single +5V supply, the output voltage can swing to within 1V
of either supply pin while delivering more than 75mA output
current. A demanding 100

load to a midpoint bias is used

in this characterization circuit. The new output stage used in
the OPA2691 can deliver large bipolar output currents into
this midpoint load with minimal crossover distortion, as shown
by the +5V supply, 3rd-harmonic distortion plots.

FIGURE 1. DC-Coupled, G = +2, Bipolar Supply, Specifica-

tion and Test Circuit.

FIGURE 2. AC-Coupled, G = +2, Single-Supply Specification

and Test Circuit.

1/2

OPA2691

+5V

DIS

Load

Source

402

6.8

0.1

6.8

0.1

0.01

1/2

OPA2691

+5V

DIS

806

100

57.6

806

453

0.1

6.8

0.1

OPA2691

SBOS224A

www.ti.com

SINGLE-SUPPLY DIFFERENTIAL ADC DRIVER

Figure 3 shows a gain of +10 Single-Ended In/Diff. Out single-
supply ADC driver. Using a dual amplifier like the OPA2691
helps reduce the necessary board space, as it also reduces the
amount of required supply bypassing components. From a
signal point of view, dual amplifiers provide excellent perfor-
mance matching (e.g., gain and phase matching). The differen-
tial ADC driver circuit shown in Figure 3 takes advantage of this
fact. A transformer converts the single-ended input signal into a
low-level differential signal which is applied to the high imped-
ance noninverting inputs of each of the two amplifiers in the
OPA2691. Resistor R

between the inverting inputs controls the

AC-gain of this circuit according to equation G = 1 + 2R

With the resistor values shown, the AC-gain is set to 10. Adding
a capacitor (0.1

F) in series with R

blocks, the DC-path gives

a DC gain of +1 for the common-mode voltage. This allows, in
a very simple way, to apply the required DC bias voltage of
+2.5V to the inputs of the amplifiers, which will also appear at
their outputs. Like the OPA2691, the ADC ADS823 operates on
a single +5V supply. Its internal common-mode voltage is
typically +2.5V which equals the required bias voltage for the
OPA2691.

Connecting two resistors between the top reference
(REFT = +3.5V) and bottom reference (REFB = +1.5V)
develops a +2.5V voltage level at their midpoint. Applying
that to the center tap of the transformer biases the ampli-
fiers appropriately. Sufficient bypassing at the center tap
must be provided to keep this point at a solid AC ground.
Resistors R

isolate the op amp output from the capacitive

input of the converter, as well as forming a 1st-order, low-
pass filter with capacitor C

to attenuate some of the

wideband noise. This interface will provide > 150MHz full-
scale input bandwidth to the ADS823.

WIDEBAND VIDEO MULTIPLEXING

One common application for video speed amplifiers which
include a disable pin is to wire multiple amplifier outputs
together, then select which one of several possible video
inputs to source onto a single line. This simple "Wired-OR
Video Multiplexer" can be easily implemented using the
OPA2691I-14D, see Figure 4.

Typically, channel switching is performed either on sync or
retrace time in the video signal. The two inputs are approxi-
mately equal at this time. The "make-before-break" disable
characteristic of the OPA2691 ensures that there is always
one amplifier controlling the line when using a wired-OR
circuit like that presented in Figure 4. Since both inputs may
be on for a short period during the transition between
channels, the outputs are combined through the output
impedance matching resistors (82.5

in this case). When

one channel is disabled, its feedback network forms part of
the output impedance and slightly attenuates the signal in
getting out onto the cable. The gain and output matching
resistors have been slightly increased to get a signal gain of
+1 at the matched load and provide a 75

output impedance

to the cable. The video multiplexer connection (see Figure 4)
also insures that the maximum differential voltage across the
inputs of the unselected channel do not exceed the rated

1.2V maximum for standard video signal levels.

The section on Disable Operation shows the turn-on and
turn-off switching glitches using a grounded input for a single
channel is typically less than

50mV. Where two outputs are

switched (see Figure 4), the output line is always under the
control of one amplifier or the other due to the "make-before-
break" disable timing. In this case, the switching glitches for
two 0V inputs drop to < 20mV.

FIGURE 3. Wideband, Single-Supply, Differential ADC Driver.

200

24.9

ADS823

10-Bit

60MSPS

24.9

0.1

GND

200

BIAS

= +2.5V

REFT

(+3.5V)

REFB

(+1.5V)

10pF

1/2

OPA2691

1/2

OPA2691

0.1

+5V

1:1

0.1

G = 1 + = 10

2 R

4.7

OPA2691

SBOS224A

www.ti.com

HIGH-SPEED ACTIVE FILTERS

Wideband current-feedback op amps make ideal elements
for implementing high-speed active filters where the amplifier
is used as a fixed gain block inside a passive RC circuit
network. Their relatively constant bandwidth versus gain,
provides low interaction between the actual filter poles and

the required gain for the amplifier. Figure 5 shows an ex-
ample single-supply buffered filter application. In this case,
one of the OPA2691 channels is used to setup the DC
operating point and provide impedance isolation from the
signal source into the 2nd-stage filter. That stage is set up to
implement a 20MHz maximally flat Butterworth frequency
response and provide an AC gain of +4.

FIGURE 5. Buffered Single-Supply Active Filter.

FIGURE 4. Two-Channel Video Multiplexer.

82.5

Cable

RG-59

82.5

402

340

Video 1

+5V

1/2

OPA2691

1/2

OPA2691

402

340

Video 2

+5V

DIS

Power-supply

decoupling not shown.

600

20MHz, 2nd-Order Butterworth Low-Pass

1/2

OPA2691

+5V

Power-supply

decoupling not shown.

32.3

105

0.1

375

1/2

OPA2691

100pF

125

0.1

150pF

Frequency (MHz)

20MHz, 2ND-ORDER BUTTERWORTH
LOW-PASS FREQUENCY RESPONSE

0.1

100

Gain (dB)

OPA2691

SBOS224A

www.ti.com

The 51

input matching resistor is optional in this case. The

input signal is AC-coupled to the 2.5V DC reference voltage
developed through the resistor divider from the +5V power
supply. This first stage acts as a gain of +1 voltage buffer for
the signal where the 600

feedback resistor is required for

stability. This first stage easily drives the low input resistors
required at the input of this high-frequency filter. The 2nd
stage is set for a DC gain of +1--carrying the 2.5V operating
point through to the output pin, and an AC gain of +4. The
feedback resistor has been adjusted to optimize bandwidth
for the amplifier itself. As the single-supply frequency re-
sponse plots show, the OPA2691 in this configuration will
give > 200MHz small-signal bandwidth. The capacitor values
were chosen as low as possible but adequate to swamp out
the parasitic input capacitance of the amplifier. The resistor
values were slightly adjusted to give the desired filter fre-
quency response while accounting for the approximate 1ns
propagation delay through each channel of the OPA2691.

HIGH-POWER TWISTED-PAIR DRIVER

A very demanding application for a high-speed amplifier is to
drive a low load impedance while maintaining a high output
voltage swing to high frequencies. Using the dual current-
feedback op amp OPA2691, a 15Vp-p output signal swing
into a twisted-pair line with a typical impedance of 100

can

be realized. Configured as shown on the front page, the two
amplifiers of the OPA2691 drive the output transformer in a
push-pull configuration thus doubling the peak-to-peak signal
swing at each op amp's output to 15Vp-p. The transformer
has a turns ratio of 2. In order to provide a matched source,
this requires a 25

source impedance (R

), for the primary

side, given the transformer equation n

= R

. Dividing this

impedance equally between the outputs requires a series
termination matching resistor at each output of 12.4

. Taking

the total resistive load of 25

(for the differential output

signal) and drawing a load line on the "Output Voltage and
Current Limitations" plot it can be seen that a 1.5V headroom
is required at both the positive and negative peak currents of
150mA.

Line driver applications usually have a high demand for
transmitting the signal with low distortion. Current-feedback
amplifiers like the OPA2691 are ideal for delivering low
distortion performance to higher gains. The example shown
is set for a differential gain of 7.5. This circuit can deliver the
maximum 15Vp-p signal with over 60MHz bandwidth.

WIDEBAND (200MHz) INSTRUMENTATION AMPLIFIER

As discussed previously, the current-feedback topology of
the OPA2691 provides a nearly constant bandwidth as signal
gain is increased. The three op amp wideband instrumenta-
tion amplifiers depicted in Figure 6 takes advantage of this,
achieving a differential bandwidth of 200MHz. The signal is
applied to the high-impedance noninverting inputs of the
OPA2691. The differential gain is set by (1 + 2R

)--which

is equal to 5 using the values shown in Figure 6. The
feedback resistors, R

, are optimized at this particular gain.

Gain adjustments can be made by adjusting R

. The differ-

ential to single-ended conversion is performed by the volt-
age-feedback amplifier OPA690, configured as a standard
difference amplifier. To maintain good distortion performance
for the OPA2691, the loading at each amplifier output has
been matched by setting R

+ R

= R

, rather than using the

same resistor values within the difference amplifier.

DEMO BOARD

ORDERING

PRODUCT

PACKAGE

NUMBER

OPA2691ID

SO-8

DEM-OPA268xU

SBOU003

OPA2691I-14D

SO-14

DEM-OPA268xN

SBOU002

261

130

249

499

261

1/2

OPA2691

1/2

OPA2691

+5V

OPA690

+5V

)

= 5

FIGURE 6. Wideband, 3-Op Amp Instrumentation Diff. Amp.

DESIGN-IN TOOLS

DEMONSTRATION BOARDS

Several PC boards are available to assist in the initial
evaluation of circuit performance using the OPA2691 in its
two package styles. Both of these are available, free, as an
unpopulated PC board delivered with descriptive documen-
tation. The summary information for these boards is shown
in the table below.

Frequency (MHz)

INSTRUMENTATION DIFF. AMP

FREQUENCY RESPONSE

400

100

Gain (dB)

To request any of these boards, check the Texas Instru-
ments web site at www.ti.com.

OPA2691

SBOS224A

www.ti.com

MACROMODELS AND APPLICATIONS SUPPORT

Computer simulation of circuit performance using SPICE is
often useful when analyzing the performance of analog
circuits and systems. This is particularly true for Video and
RF amplifier circuits where parasitic capacitance and induc-
tance can have a major effect on circuit performance. A
SPICE model for the OPA2691 is available through the TI
web site (www.ti.com). These models do a good job of
predicting small-signal AC and transient performance under
a wide variety of operating conditions. They do not do as well
in predicting the harmonic distortion or dG/dP characteristics.
These models do not attempt to distinguish between the
package types in their small-signal AC performance, nor do
they attempt to simulate channel-to-channel coupling.

OPERATING SUGGESTIONS

SETTING RESISTOR VALUES TO
OPTIMIZE BANDWIDTH

A current-feedback op amp like the OPA2691 can hold an
almost constant bandwidth over signal gain settings with the
proper adjustment of the external resistor values. This is
shown in the Typical Characteristics; the small-signal band-
width decreases only slightly with increasing gain. Those
curves also show that the feedback resistor has been changed
for each gain setting. The resistor "values" on the inverting
side of the circuit for a current-feedback op amp can be
treated as frequency response compensation elements while
their "ratios" set the signal gain. Figure 7 shows the small-
signal frequency response analysis circuit for the OPA2691.

, the buffer output impedance, is a critical portion of the

bandwidth control equation. The OPA2691 is typically about 37

A current-feedback op amp senses an error current in the
inverting node (as opposed to a differential input error volt-
age for a voltage-feedback op amp) and passes this on to the
output through an internal frequency-dependent transimped-
ance gain. The Typical Characteristics show this open-loop
transimpedance response. This is analogous to the open-
loop voltage gain curve for a voltage-feedback op amp.
Developing the transfer function for the circuit of Figure 7
gives Equation 1:

R NG

( )

(1)

This is written in a loop-gain analysis format where the errors
arising from a non-infinite open-loop gain are shown in the
denominator. If Z

(S)

were infinite over all frequencies, the

denominator of Equation 1 would reduce to 1 and the ideal
desired signal gain shown in the numerator would be achieved.
The fraction in the denominator of Equation 1 determines the
frequency response. Equation 2 shows this as the loop-gain
equation:

R NG

Loop Gain

( )

(2)

If 20 · log(R

+ NG · R

) were drawn on top of the open-loop

transimpedance plot, the difference between the two would
be the loop gain at a given frequency. Eventually, Z

(S)

rolls off

to equal the denominator of Equation 2 at which point the
loop gain has reduced to 1 (and the curves have intersected).
This point of equality is where the amplifier's closed-loop
frequency response given by Equation 1 will start to roll off,
and is exactly analogous to the frequency at which the noise
gain equals the open-loop voltage gain for a voltage-feed-
back op amp. The difference here is that the total impedance
in the denominator of Equation 2 may be controlled some-
what separately from the desired signal gain (or NG).

The OPA2691 is internally compensated to give a maximally
flat frequency response for R

= 402

at NG = 2 on

supplies. Evaluating the denominator of Equation 2 (which is
the feedback transimpedance) gives an optimal target of 476

As the signal gain changes, the contribution of the NG · R

term

in the feedback transimpedance will change, but the total can
be held constant by adjusting R

. Equation 3 gives an approxi-

mate equation for optimum R

over signal gain:

NG R

476

(3)

As the desired signal gain increases, this equation will even-
tually predict a negative R

. A somewhat subjective limit to this

adjustment can also be set by holding R

to a minimum value

of 20

. Lower values will load both the buffer stage at the input

and the output stage if R

gets too low--actually decreasing

The key elements of this current-feedback op amp model are:

Buffer gain from the noninverting input to the inverting input

Buffer output impedance

ERR

Feedback error current signal

Z(s)

Frequency dependent open-loop transimpedance

gain from i

ERR

to V

The buffer gain is typically very close to 1.00 and is normally
neglected from signal gain considerations. It will, however, set
the CMRR for a single op amp differential amplifier configura-
tion. For a buffer gain

< 1.0, the CMRR = 20 · log (1

)dB.

FIGURE 7. Current-Feedback Transfer Function Analysis Circuit.

(S)

ERR

OPA2691

SBOS224A

www.ti.com

the bandwidth. Figure 8 shows the recommended R

versus

NG for both

5V and a single +5V operation. The values for R

versus Gain shown here are approximately equal to the values
used to generate the Typical Characteristics. They differ in that
the optimized values used in the Typical Characteristics are
also correcting for board parasitics not considered in the
simplified analysis leading to Equation 3. The values shown in
Figure 8 give a good starting point for design where bandwidth
optimization is desired.

impedance matching is desired (which is beneficial when-
ever the signal is coupled through a cable, twisted-pair, long
PC board trace, or other transmission line conductor), it is
normally necessary to add an additional matching resistor to
ground. R

by itself is normally not set to the required input

impedance since its value, along with the desired gain, will
determine an R

which may be non-optimal from a frequency

response standpoint. The total input impedance for the
source becomes the parallel combination of R

and R

The second major consideration, touched on in the previous
paragraph, is that the signal source impedance becomes
part of the noise gain equation and will have slight effect on
the bandwidth through Equation 1. The values shown in
Figure 9 have accounted for this by slightly decreasing R

(from Figure 1) to re-optimize the bandwidth for the noise
gain of Figure 9 (NG = 2.74) In the example of Figure 9, the
R

value combines in parallel with the external 50

source

impedance, yielding an effective driving impedance of
50

|| 68

= 28.8

. This impedance is added in series with

for calculating the noise gain--which gives NG = 2.74.

This value, along with the R

of Figure 8 and the inverting

input impedance of 37

, are inserted into Equation 3 to get

a feedback transimpedance nearly equal to the 476

opti-

mum value.

Note that the noninverting input in this bipolar supply invert-
ing application is connected directly to ground. It is often
suggested that an additional resistor be connected to ground
on the noninverting input to achieve bias current error can-
cellation at the output. The input bias currents for a current-
feedback op amp are not generally matched in either magni-
tude or polarity. Connecting a resistor to ground on the
noninverting input of the OPA2691 in the circuit of Figure 9
will actually provide additional gain for that input's bias and
noise currents, but will not decrease the output DC error
since the input bias currents are not matched.

OUTPUT CURRENT AND VOLTAGE

The OPA2691 provides output voltage and current capabili-
ties that are unsurpassed in a low-cost, dual monolithic op
amp. Under no-load conditions at 25

C, the output voltage

typically swings closer than 1V to either supply rail; the tested

600

500

400

300

200

100

Noise Gain

Feedback Resistor (

)

+5V

FIGURE 8. Recommended Feedback Resistor vs Noise Gain.

1/2

OPA2691

374

182

+5V

Load

Power-supply

decoupling not shown.

Source

68.1

FIGURE 9. Inverting Gain of 2 with Impedance Matching.

The total impedance going into the inverting input may be
used to adjust the closed-loop signal bandwidth. Inserting a
series resistor between the inverting input and the summing
junction will increase the feedback impedance (denominator
of Equation 2), decreasing the bandwidth. The internal buffer
output impedance for the OPA2691 is slightly influenced by
the source impedance looking out of the noninverting input
terminal. High source resistors will have the effect of increas-
ing R

, decreasing the bandwidth. For those single-supply

applications which develop a midpoint bias at the noninverting
input through high valued resistors, the decoupling capacitor
is essential for power-supply ripple rejection, noninverting
input noise current shunting, and to minimize the high-
frequency value for R

in Figure 7.

INVERTING AMPLIFIER OPERATION

Since the OPA2691 is a general-purpose, wideband current-
feedback op amp, most of the familiar op amp application
circuits are available to the designer. Those dual op amp
applications that require considerable flexibility in the feed-
back element (e.g., integrators, transimpedance, and some
filters) should consider the unity-gain stable voltage-feed-
back OPA2690, since the feedback resistor is the compen-
sation element for a current-feedback op amp. Wideband
inverting operation (and especially summing) is particularly
suited to the OPA2691. Figure 9 shows a typical inverting
configuration where the I/O impedances and signal gain from
Figure 1 are retained in an inverting circuit configuration.

In the inverting configuration, two key design considerations
must be noted. The first is that the gain resistor (R

)

becomes part of the signal channel input impedance. If input

OPA2691

SBOS224A

www.ti.com

swing limit is within 1.2V of either rail. Into a 15

load (the

minimum tested load), it is tested to deliver more than

160mA.

The specifications described above, though familiar in the
industry, consider voltage and current limits separately. In
many applications, it is the voltage x current, or V-I product,
which is more relevant to circuit operation. Refer to the
"Output Voltage and Current Limitations" plot in the Typical
Characteristics. The X- and Y-axes of this graph show the
zero-voltage output current limit and the zero-current output
voltage limit, respectively. The four quadrants give a more
detailed view of the OPA2691's output drive capabilities,
noting that the graph is bounded by a "Safe Operating Area"
of 1W maximum internal power dissipation (in this case for 1
channel only). Superimposing resistor load lines onto the plot
shows that the OPA2691 can drive

2.5V into 25

3.5V

into 50

without exceeding the output capabilities or the 1W

dissipation limit. A 100

load line (the standard test circuit

load) shows the full

3.9V output swing capability, as shown

in the Electrical Characteristics.

The minimum specified output voltage and current over
temperature are set by worst-case simulations at the cold
temperature extreme. Only at cold start-up will the output
current and voltage decrease to the numbers shown in the
electrical characteristic tables. As the output transistors de-
liver power, their junction temperatures will increase, de-
creasing their V

's (increasing the available output voltage

swing) and increasing their current gains (increasing the
available output current). In steady-state operation, the avail-
able output voltage and current will always be greater than
that shown in the over-temperature specifications since the
output stage junction temperatures will be higher than the
minimum specified operating ambient.

To protect the output stage from accidental shorts to ground
and the power supplies, output short-circuit protection is
included in the OPA2691. The circuit acts to limit the maxi-
mum source or sink current to approximately 250mA.

DRIVING CAPACITIVE LOADS

One of the most demanding and yet very common load
conditions for an op amp is capacitive loading. Often, the
capacitive load is the input of an ADC--including additional
external capacitance which may be recommended to im-
prove the ADC's linearity. A high-speed, high open-loop gain
amplifier like the OPA2691 can be very susceptible to de-
creased stability and closed-loop response peaking when a
capacitive load is placed directly on the output pin. When the
amplifier's open-loop output resistance is considered, this
capacitive load introduces an additional pole in the signal
path that can decrease the phase margin. Several external
solutions to this problem have been suggested. When the
primary considerations are frequency response flatness, pulse
response fidelity, and/or distortion, the simplest and most
effective solution is to isolate the capacitive load from the
feedback loop by inserting a series isolation resistor between
the amplifier output and the capacitive load. This does not
eliminate the pole from the loop response, but rather shifts it

and adds a zero at a higher frequency. The additional zero
acts to cancel the phase lag from the capacitive load pole,
thus increasing the phase margin and improving stability.

The Typical Characteristics show the recommended "R

Capacitive Load" and the resulting frequency response at the
load. Parasitic capacitive loads greater than 2pF can begin to
degrade the performance of the OPA2691. Long PC board
traces, unmatched cables, and connections to multiple de-
vices can easily cause this value to be exceeded. Always
consider this effect carefully, and add the recommended
series resistor as close as possible to the OPA2691 output
pin (see Board Layout Guidelines).

DISTORTION PERFORMANCE

The OPA2691 provides good distortion performance into a
100

load on

5V supplies. Relative to alternative solutions,

it provides exceptional performance into lighter loads and/or
operating on a single +5V supply. Generally, until the funda-
mental signal reaches very high frequency or power levels,
the 2nd-harmonic will dominate the distortion with a negligible
3rd-harmonic component. Focusing then on the 2nd-har-
monic, increasing the load impedance improves distortion
directly. Remember that the total load includes the feedback
network--in the noninverting configuration (see Figure 1) this
is the sum of R

+ R

, while in the inverting configuration it is

just R

. Also, providing an additional supply decoupling ca-

pacitor (0.01

F) between the supply pins (for bipolar opera-

tion) improves the 2nd-order distortion slightly (3dB to 6dB).

In most op amps, increasing the output voltage swing in-
creases harmonic distortion directly. The Typical Character-
istics show the 2nd-harmonic increasing at a little less than
the expected 2x rate while the 3rd-harmonic increases at a
little less than the expected 3x rate. Where the test power
doubles, the difference between it and the 2nd-harmonic
decreases less than the expected 6dB while the difference
between it and the 3rd-harmonic decreases by less than the
expected 12dB. This also shows up in the 2-tone, 3rd-order
intermodulation spurious (IM3) response curves. The 3rd-
order spurious levels are extremely low at low output power
levels. The output stage continues to hold them low even as
the fundamental power reaches very high levels. As the
Typical Characteristics show, the spurious intermodulation
powers do not increase as predicted by a traditional intercept
model. As the fundamental power level increases, the dy-
namic range does not decrease significantly. For two tones
centered at 20MHz, with 10dBm/tone into a matched 50

load (i.e., 2Vp-p for each tone at the load, which requires
8Vp-p for the overall 2-tone envelope at the output pin), the
Typical Characteristics show 48dBc difference between the
test-tone power and the 3rd-order intermodulation spurious
levels. This exceptional performance improves further when
operating at lower frequencies.

NOISE PERFORMANCE

Wideband current-feedback op amps generally have a higher
output noise than comparable voltage-feedback op amps. The
OPA2691 offers an excellent balance between voltage and

OPA2691

SBOS224A

www.ti.com

current noise terms to achieve low output noise. The inverting
current noise (15pA/

Hz) is significantly lower than earlier

solutions while the input voltage noise (1.7nV/

Hz) is lower

than most unity-gain stable, wideband, voltage-feedback op
amps. This low input voltage noise was achieved at the price
of higher noninverting input current noise (12pA/

Hz). As long

as the AC source impedance looking out of the noninverting
node is less than 100

, this current noise will not contribute

significantly to the total output noise. The op amp input voltage
noise and the two input current noise terms combine to give
low output noise under a wide variety of operating conditions.
Figure 10 shows the op amp noise analysis model with all the
noise terms included. In this model, all noise terms are taken
to be noise voltage or current density terms in either nV/

or pA/

Hz.

DC ACCURACY AND OFFSET CONTROL

A current-feedback op amp like the OPA2691 provides
exceptional bandwidth in high gains, giving fast pulse settling
but only moderate DC accuracy. The Electrical Characteris-
tics show an input offset voltage comparable to high-speed
voltage-feedback amplifiers. However, the two input bias
currents are somewhat higher and are unmatched. Whereas
bias current cancellation techniques are very effective with
most voltage-feedback op amps, they do not generally re-
duce the output DC offset for wideband current-feedback op
amps. Since the two input bias currents are unrelated in both
magnitude and polarity, matching the source impedance
looking out of each input to reduce their error contribution to
the output is ineffective. Evaluating the configuration of
Figure 1, using worst-case +25

C input offset voltage and the

two input bias currents, gives a worst-case output offset
range equal to:

(NG · V

OS(MAX)

) + (I

· R

/2 · NG)

· R

)

where NG = noninverting signal gain

(2 · 3.0mV) + (35

A · 25

· 2)

(402

· 25

6mV + 1.75mV

10.05mV

= 14.3mV

+17.8mV

DISABLE OPERATION (SO-14 ONLY)

The OPA2691I-14D provides an optional disable feature that
may be used either to reduce system power or to implement
a simple channel multiplexing operation. If the DIS control
pin is left unconnected, the OPA2691I-14D will operate
normally. To disable, the control pin must be asserted low.
Figure 11 shows a simplified internal circuit for the disable
control feature.

The total output spot noise voltage can be computed as the
square root of the sum of all squared output noise voltage
contributors. Equation 4 shows the general form for the
output noise voltage using the terms shown in Figure 10.

(4)

kTR NG

I R

kTR NG

(

)

(

)

(

)

Dividing this expression by the noise gain (NG = (1 + R

))

will give the equivalent input referred spot noise voltage at
the noninverting input, as shown in Equation 5.

kTR

I R

kTR

(

)

(5)

Evaluating these two equations for the OPA2691 circuit and
component values presented in Figure 1 will give a total
output spot noise voltage of 8.08nV/

Hz and a total equiva-

lent input spot noise voltage of 4.04nV/

Hz. This total input

referred spot noise voltage is higher than the 1.7nV/

specification for the op amp voltage noise alone. This reflects
the noise added to the output by the inverting current noise
times the feedback resistor. If the feedback resistor is re-
duced in high-gain configurations (as suggested previously),
the total input referred voltage noise given by Equation 5 will
approach just the 1.7nV/

Hz of the op amp itself. For

example, going to a gain of +10 using R

= 180

will give a

total input referred noise of 2.1nV/

Hz.

In normal operation, base current to Q1 is provided through
the 110k

resistor while the emitter current through the 15k

resistor sets up a voltage drop that is inadequate to turn on
the two diodes in Q1's emitter. As V

DIS

is pulled low, addi-

tional current is pulled through the 15k

resistor, eventually

turning on these two diodes (

100

A). At this point, any

further current pulled out of V

DIS

goes through those diodes

FIGURE 10. Op Amp Noise Analysis Model.

4kT

1/2

OPA2691

4kT = 1.6E 20J

at 290

4kTR

25k

110k

15k

Control

DIS

FIGURE 11. Simplified Disable Control Circuit, Each Channel.

OPA2691

SBOS224A

www.ti.com

THERMAL ANALYSIS

Due to the high output power capability of the OPA2691,
heatsinking or forced airflow may be required under extreme
operating conditions. Maximum desired junction temperature
will set the maximum allowed internal power dissipation as
described below. In no case should the maximum junction
temperature be allowed to exceed 175

C. Operating junction

temperature (T

) is given by T

+ P

. The total internal

power dissipation (P

) is the sum of quiescent power (P

)

and additional power dissipation in the output stage (P

) to

deliver load power. Quiescent power is simply the specified
no-load supply current times the total supply voltage across
the part. P

will depend on the required output signal and

load but would, for a grounded resistive load, be at a
maximum when the output is fixed at a voltage equal to 1/2
of either supply voltage (for equal bipolar supplies). Under
this condition, P

= V

/(4 · R

) where R

includes feedback

network loading.

Note that it is the power in the output stage and not into the
load that determines internal power dissipation.

As a worst-case example, compute the maximum T

using an

OPA2691 SO-8 in the circuit of Figure 1 operating at the
maximum specified ambient temperature of +85

C with both

outputs driving a grounded 20

load to +2.5V.

= 10V · 11.5mA + 2 · [5

/(4 · (20

|| 804

))] = 756mW

Maximum T

= +85

C + (0.756 · 125

C/W) = 179.5

This absolute worst-case condition exceeds specified maxi-
mum junction temperature. Normally this extreme case will
not be encountered. Careful attention to internal power
dissipation is required.

BOARD LAYOUT GUIDELINES

Achieving optimum performance with a high-frequency ampli-
fier like the OPA2691 requires careful attention to board
layout parasitics and external component types. Recommen-
dations that will optimize performance include:

a) Minimize parasitic capacitance to any AC ground for all
of the signal I/O pins. Parasitic capacitance on the output and
inverting input pins can cause instability: on the noninverting
input, it can react with the source impedance to cause
unintentional bandlimiting. To reduce unwanted capacitance,
a window around the signal I/O pins should be opened in all
of the ground and power planes around those pins. Other-
wise, ground and power planes should be unbroken else-
where on the board.

b) Minimize the distance (< 0.25") from the power-supply
pins to high-frequency 0.1

F decoupling capacitors. At the

device pins, the ground and power plane layout should not
be in close proximity to the signal I/O pins. Avoid narrow
power and ground traces to minimize inductance between
the pins and the decoupling capacitors. The power-supply
connections (on pins 4 and 7) should always be decoupled
with these capacitors. An optional supply decoupling capaci-
tor across the two power supplies (for bipolar operation) will
improve 2nd-harmonic distortion performance. Larger (2.2

to 6.8

F) decoupling capacitors, effective at lower frequency,

should also be used on the main supply pins. These may be
placed somewhat farther from the device and may be shared
among several devices in the same area of the PC board.

c) Careful selection and placement of external compo-
nents will preserve the high-frequency performance of
the OPA2691. Resistors should be a very low reactance
type. Surface-mount resistors work best and allow a tighter
overall layout. Metal film and carbon composition axially-
leaded resistors can also provide good high-frequency per-
formance. Again, keep their leads and PC board trace length

holding the emitter-base voltage of Q1 at approximately 0V.
This shuts off the collector current out of Q1, turning the
amplifier off. The supply currents in the disable mode are only
those required to operate the circuit of Figure 11. Additional
circuitry ensures that turn-on time occurs faster than turn-off
time (make-before-break).

When disabled, the output and input nodes go to a high
impedance state. If the OPA2691 is operating in a gain of +1,
this will show a very high impedance (4pF || 1M

) at the

output and exceptional signal isolation. If operating at a
gain greater than +1, the total feedback network resistance
(R

+ R

) will appear as the impedance looking back into the

output, but the circuit will still show very high forward and
reverse isolation. If configured as an inverting amplifier, the
input and output will be connected through the feedback
network resistance (R

+ R

) giving relatively poor input to

output isolation.

One key parameter in disable operation is the output glitch
when switching in and out of the disabled mode. Figure 12
shows these glitches for the circuit of Figure 1 with the input
signal set to 0V. The glitch waveform at the output pin is
plotted along with the DIS pin voltage.

The transition edge rate (dv/dt) of the DIS control line will
influence this glitch. For the plot of Figure 12, the edge rate
was reduced until no further reduction in glitch amplitude was
observed. This approximately 1V/ns maximum slew rate may
be achieved by adding a simple RC filter into the V

DIS

from a higher speed logic line. If extremely fast transition
logic is used, a 2k

series resistor between the logic gate

and the V

DIS

input pin will provide adequate bandlimiting

using just the parasitic input capacitance on the V

DIS

while still ensuring adequate logic level swing.

FIGURE 12. Disable/Enable Glitch.

Time (20ns/div)

Output Voltage (10mV/div)

6.0

4.0

2.0

0.0

DIS

(2V/div)

OPA2691

SBOS224A

www.ti.com

as short as possible. Never use wirewound type resistors in
a high-frequency application. Since the output pin and invert-
ing input pin are the most sensitive to parasitic capacitance,
always position the feedback and series output resistor, if
any, as close as possible to the output pin. Other network
components, such as noninverting input termination resis-
tors, should also be placed close to the package. Where
double-side component mounting is allowed, place the feed-
back resistor directly under the package on the other side of
the board between the output and inverting input pins. The
frequency response is primarily determined by the feedback
resistor value as described previously. Increasing its value
will reduce the bandwidth, while decreasing it will give a more
peaked frequency response. The 402

feedback resistor

used in the electrical characteristics at a gain of +2 on

supplies is a good starting point for design. Note that a 453

feedback resistor, rather than a direct short, is recommended
for the unity-gain follower application. A current-feedback op
amp requires a feedback resistor even in the unity-gain
follower configuration to control stability.

d) Connections to other wideband devices on the board
may be made with short direct traces or through onboard
transmission lines. For short connections, consider the trace
and the input to the next device as a lumped capacitive load.
Relatively wide traces (50mils to 100mils) should be used,
preferably with ground and power planes opened up around
them. Estimate the total capacitive load and set R

from the

plot of "Recommended R

vs Capacitive Load". Low parasitic

capacitive loads (< 5pF) may not need an R

since the

OPA2691 is nominally compensated to operate with a 2pF
parasitic load. If a long trace is required, and the 6dB signal
loss intrinsic to a doubly-terminated transmission line is
acceptable, implement a matched impedance transmission
line using microstrip or stripline techniques (consult an ECL
design handbook for microstrip and stripline layout tech-
niques). A 50

environment is normally not necessary on

board, and in fact a higher impedance environment will
improve distortion as shown in the "Distortion vs Load" plots.
With a characteristic board trace impedance defined based
on board material and trace dimensions, a matching series
resistor into the trace from the output of the OPA2691 is used
as well as a terminating shunt resistor at the input of the
destination device. Remember also that the terminating im-
pedance will be the parallel combination of the shunt resistor
and the input impedance of the destination device: this total
effective impedance should be set to match the trace imped-
ance. The high output voltage and current capability of the
OPA2691 allows multiple destination devices to be handled

as separate transmission lines, each with their own series
and shunt terminations. If the 6dB attenuation of a doubly-
terminated transmission line is unacceptable, a long trace
can be series-terminated at the source end only. Treat the
trace as a capacitive load in this case and set the series
resistor value as shown in the plot of "R

vs Capacitive

Load". This will not preserve signal integrity as well as a
doubly-terminated line. If the input impedance of the destina-
tion device is low, there will be some signal attenuation due
to the voltage divider formed by the series output into the
terminating impedance.

e) Socketing a high-speed part like the OPA2691 is not
recommended. The additional lead length and pin-to-pin
capacitance introduced by the socket can create an ex-
tremely troublesome parasitic network which can make it
almost impossible to achieve a smooth, stable frequency
response. Best results are obtained by soldering the OPA2691
onto the board.

INPUT AND ESD PROTECTION

The OPA2691 is built using a very high-speed complemen-
tary bipolar process. The internal junction breakdown volt-
ages are relatively low for these very small geometry de-
vices. These breakdowns are reflected in the Absolute Maxi-
mum Ratings table. All device pins have limited ESD protec-
tion using internal diodes to the power supplies, as shown in
Figure 13.

These diodes provide moderate protection to input overdrive
voltages above the supplies as well. The protection diodes
can typically support 30mA continuous current. Where higher
currents are possible (e.g., in systems with

15V supply parts

driving into the OPA2691), current-limiting series resistors
should be added into the two inputs. Keep these resistor
values as low as possible since high values degrade both
noise performance and frequency response.

FIGURE 12. Internal ESD Protection.

External

Internal
Circuitry

IMPORTANT NOTICE

Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications,
enhancements, improvements, and other changes to its products and services at any time and to discontinue
any product or service without notice. Customers should obtain the latest relevant information before placing
orders and should verify that such information is current and complete. All products are sold subject to TI's terms
and conditions of sale supplied at the time of order acknowledgment.

TI warrants performance of its hardware products to the specifications applicable at the time of sale in
accordance with TI's standard warranty. Testing and other quality control techniques are used to the extent TI
deems necessary to support this warranty. Except where mandated by government requirements, testing of all
parameters of each product is not necessarily performed.

TI assumes no liability for applications assistance or customer product design. Customers are responsible for
their products and applications using TI components. To minimize the risks associated with customer products
and applications, customers should provide adequate design and operating safeguards.

TI does not warrant or represent that any license, either express or implied, is granted under any TI patent right,
copyright, mask work right, or other TI intellectual property right relating to any combination, machine, or process
in which TI products or services are used. Information published by TI regarding third-party products or services
does not constitute a license from TI to use such products or services or a warranty or endorsement thereof.
Use of such information may require a license from a third party under the patents or other intellectual property
of the third party, or a license from TI under the patents or other intellectual property of TI.

Reproduction of information in TI data books or data sheets is permissible only if reproduction is without
alteration and is accompanied by all associated warranties, conditions, limitations, and notices. Reproduction
of this information with alteration is an unfair and deceptive business practice. TI is not responsible or liable for
such altered documentation.

Resale of TI products or services with statements different from or beyond the parameters stated by TI for that
product or service voids all express and any implied warranties for the associated TI product or service and
is an unfair and deceptive business practice. TI is not responsible or liable for any such statements.

Following are URLs where you can obtain information on other Texas Instruments products and application
solutions:

Products

Applications

Amplifiers

amplifier.ti.com

Audio

www.ti.com/audio

Data Converters

dataconverter.ti.com

Automotive

www.ti.com/automotive

DSP

dsp.ti.com

Broadband

www.ti.com/broadband

Interface

interface.ti.com

Digital Control

www.ti.com/digitalcontrol

Logic

logic.ti.com

Military

www.ti.com/military

Power Mgmt

power.ti.com

Optical Networking

www.ti.com/opticalnetwork

Microcontrollers

microcontroller.ti.com

Security

www.ti.com/security

Telephony

www.ti.com/telephony

Video & Imaging

www.ti.com/video

Wireless

www.ti.com/wireless

Mailing Address:

Texas Instruments

Post Office Box 655303 Dallas, Texas 75265

2003, Texas Instruments Incorporated

Ð­Ð»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: OPA2691I-14DR

ÐÐ»ÐµÐºÑ‚Ñ€Ð¾Ð½Ð½Ñ‹Ð¹ ÐºÐ¾Ð¼Ð¿Ð¾Ð½ÐµÐ½Ñ‚: OPA2691I-14DR